Transmit/Receive Module for L-Band Phased Array

Synthetic Aperture Radar with Polarisation Control

Thomas Cooney, Torsten Lehmann, Kevin Parkinson,

Robert Middleton

Australian Center for Space Engineering Research

University of New South Wales

Sydney, Australia

thomas.cooney@unswalumni.com

Charles D. Hall

ENS Division

Astrium Limited

Portsmouth, United Kingdom

david.hall@astrium.eads.net

Abstract—This paper describes the design, construction and

testing of a front end Transmit/Receive (T/R) module prototype

for an L-band (1.215GHz-1.3GHz) synthetic aperture radar

earth observation satellite which features a novel polarisation

control circuit. This circuit is implemented to support the on-

board Faraday Rotation compensation. Previous literature does

not feature this control at L-band or in a single T/R module. The

prototype consists of Monolithic Microwave Integrated Circuits

mounted on a Printed Circuit Board, and laboratory testing

shows the polarisation control mechanism functions as designed,

as measured results closely match predictions.

Keywords—Transmit/Receive, T/R, TR, Polarisation Control,

SAR, Phased Array

I. INTRODUCTION

An L-band (1.215GHz-1.3GHz) synthetic aperture radar (SAR) earth observation satellite would be a powerful soil moisture monitoring tool [1] to assist with irrigation management in the Murray-Darling basin. The mission drives the band selection, as L-band provides sufficient foliage penetration and biomass saturation sensitivity. This paper describes the design, construction and testing of a front end Transmit/Receive (T/R) module prototype for this satellite, which features a novel polarisation control circuit.

A problem that confronts L-band satellite radar imaging is Faraday Rotation (FR), which is a rotation of the angle of polarisation of Electro-Magnetic Radiation (EMR) when passing through magneto-optical media in the presence of a magnetic field [2]. The Earth’s ionosphere provides such a magneto-optical medium, rotating the polarisation of transmitted radar pulses and their received echo. In order for polarimetric channels to provide meaningful measurements in HH (Horizontal, Horizontal), VV (Vertical, Vertical), HV and VH channels, it is necessary to correct for these rotations.

T/R modules are critical components of Active Electronically Scanned Array Radars (AESARs) [3]. They are the front-end module that connects directly to a single element of the antenna array, or to several elements arranged in a subarray. They must provide power and phase/amplitude control to the signal to be transmitted and the signal to be received. Fig. 1 shows a typical structure of a T/R module.

High Power

Amplifier

Attenuator with

Digital Control

Phase Shifter with

Digital Control

Circulator

/Switch

Circulator

/Switch

From/To Array

Feed Network

To/From Antenna

Element

Phase Shifter with

Digital Control

Low Noise

Amplifier

Attenuator with

Digital Control

Figure 1. Typical T/R Module Block Diagram [4]

Tx/Rx functionality is time multiplexed; the multiplexing is provided by switches/circulators at the antenna and array feed. In Tx mode an RF signal is input to the module from the array feed network, passed through the digitally controlled phase shifter and attenuator for array beamforming before going through a High Power Amplifier (HPA) and sent to the antenna. In Rx mode, signals from the antenna are amplified by a Low Noise Amplifier (LNA) before passing through the digitally controlled phase shifter and attenuator, again for array beamforming.

Several designs from literature were considered to inform the design of the prototype. Three of the most relevant are detailed here. First is an L-Band SAR T/R module implemented by Jet Propulsion Laboratories (JPL) [5]. The module can transmit in either V or H, and receive in both V and H simultaneously. The transmit chain has 42dB of gain on a 10dBm signal from the corporate RF feed, and the receive chain has 25dB-30dB of gain. The module’s phase shifter has a 7 bit resolution. The components are all plastic surface mount Monolithic Microwave Integrated Circuits (MMICs), interconnected with microstrip transmission lines, and not qualified for space flight. JPL has also developed T/R modules in L-band for Large Aperture Membrane Arrays [6], [7] and has done significant work in the area of module self-calibration [8].

Second is an X band (9.5GHz-9.8GHz) Standard Modular T/R (SMTR) module developed and manufactured by EADS Deutschland for use on TerraSAR-X and other radar systems. The SMTR module consists of MMICs custom manufactured by United Monolithic Semiconductors and packaged together

using low temperature co-fired ceramic (LTCC) technology [9]. It can generate a transmit power of 38dBm.

Third is a T/R module by the Beijing Institute of Technology [10], which implemented polarisation control. This scheme requires a subarray with 4 panels, each with a single feed. By configuring the relative phase and gain between these feeds, an arbitrary polarisation can be generated in the far field. This approach requires 4 separate T/R modules, one for each panel.

Given the absence of single T/R module electronic control of the transmit polarisation in L-band, we will implement this novel feature in the prototype described in this paper. This feature is required to support the implementation of patent [11] on the satellite so it can perform on-board FR compensation, thereby mitigating the effects of Faraday rotation on imaging.

As in the JPL design, a Microwave Monolithic Integrated Circuit (MMIC) on PCB concept is chosen for implementation in this work. The MMICs used are Commercially available Off The Shelf (COTS) components to minimise design cost, as datasheets and application notes are readily available from manufacturers and devices are provided in standard packaging. This approach means a wide range of components designed for telecommunications that work in the 1.215GHz-1.3GHz band are available. Design of Application Specific Integrated Circuits (ASICs) for the prototype was considered as per the Astrium SMTR design, but the advantage of this approach in terms of customisation would not be fully realised in a prototype.

II. DESIGN

A. System Requirements

System requirements are set by the Garada project team. The module is to support simultaneous H and V polarisation reception, and 360° transmit polarisation control. The module must interface with an annular slot antenna with two orthogonal ports, each port providing balanced stimulation from a single RF source. The ports are arranged such that stimulation on one port will produce radiation with a polarisation angle of 45° w.r.t horizontal, and the other will produce -45° w.r.t horizontal.

360° independent phase control is required for the transmit path and each receive polarisation path. Input power is provided from a single array feed at 10mW. Both the array feed and antenna ports will have at least 15dB return loss as seen from the module. In addition, a calibration path is required to measure the total gain and phase distortion introduced from the transmit and receive path in series. The required transmit output power is 16W at the desired polarisation, and required amplification for each receive polarisation path is 30dB with a noise figure of less than 2.5dB. Pulse repeat frequency is to be no more than 2450Hz.

B. Polarisation Control

To control the polarisation of radiated EMR from the annular slot antenna, we propose the circuit shown in Fig. 2.

A

Aλ/4

λ/4

λ/4

λ/4

vtb

vta v+45

v-45

Z0

90° Hybrid Coupler 1

Phase

Shifter +β

Phase

Shifter +α HPA 1

90° Hybrid Coupler 2

To AntennaFrom Array

HPA 2

Figure 2. Polarisation Control Circuit

A is the gain of the High Power Amplifiers (HPAs), α and β are the phase advances of their respective phase shifters, vta = Re

i0 and vtb = 0 are the input voltages (R is the input voltage

magnitude), and vta and vta are the output voltages that feed into the orthogonal antenna ports. This circuit operates the HPAs at a constant power, and uses α and β to control the amplitude of the signal at the output ports of 90° Hybrid Coupler 2. Using ideal transmission coefficients for each component we derive the transfer function in equation (2).

01

1

2

1

0

0

1

1

2

1 0

45

45i

i

i eR

i

i

e

eA

i

i

v

v

(1)

2cos

2sin

2

45

45

i

ARv

v (2)

This transfer function shows there are 4 possible sign combinations of the phasor representation of the two signals, ++,--, +-, and -+. These states determine the phase difference between the two antenna feeds, and therefore determine the quadrant the polarisation angle will lie in. The relative magnitudes of the signals determine the polarisation angle within this quadrant. Both these quantities are determined by the difference in the phase advance (α - β). The polarisation angle and phase centre of the transmit circuit based on this transfer function is plotted in Fig. 3.

Figure 3. Polarisation Angle and Phase Centre

This model confirms that this circuit can provide 360° polarisation control, however the phase centre for each polarisation setting does not have a 360° range. To achieve this, an additional phase shifter must be placed before the polarisation control circuit.

Figure 4. System Block Diagram

C. System Design

50Ω characteristic impedance components are selected to maximise compatibility and the range of components available. Components are also selected on vendor availability and lead time. A system block diagram is presented in Fig. 4.

To meet the requirement of 10mW in and 16W out in the transmit path, a net gain of 32dB is required. This corresponds to A in equation (2), and is the gain of each amplifier in the polarisation control circuit of Fig. 2. The elements of the transmit path have nominal total insertion loss of 16.155dB.

Therefore the target for A is 49dB for a slight margin of 0.845dB. This is provided by two driver amplifiers with 17.5dB and 18dB gain respectively, and a final power amplifier stage of 13.5dB gain. Each receive path has 13.11dB nominal total insertion loss, and the two Low Noise Amplifiers (LNAs) selected have a total gain of 48dB, satisfying the requirement of 30dB gain.

For the calibration path from the transmit output to the receive input, the attenuator value must be appropriately selected to ensure that the power entering HG LNA 1 and 2 from the calibration path is 40dB higher than the power entering HG LNA 1 and 2 from the antenna when switched to calibration mode (an upper limit), and that the output of the LNA should be 5dB less than the 1dB compression point to permit a linear response (a lower limit). The attenuation value is chosen as 77dB, which is within these limits.

Digital control of the phase shifters, attenuators, and switches in the prototype is performed by an FPGA that is connected to the module and directly controls each component digital input. A simple interface is designed to permit individual control of each digital input using the switches and pushbuttons provided on the FPGA board. This simple, direct interface greatly simplifies testing.

Test accesses to the RF path are provided around many components on the board. These accesses consist of a mechanical switch that activates when the access is mated to a probe, providing in-place measurement of signals. They were of great benefit during testing.

D. PCB Design and Manufacture

A 4 layer PCB concept is selected. RF signals travel on the top layer, and are isolated from the supply voltage layer and control signal/thermal dissipation layer by a ground plane. 50Ω transmission lines are implemented as microstrip lines on the RF plane and surrounded by a grounded copper flood; the

Figure 4. System Block Diagram

-45°Rx/Cal Feed

Network

Tx/+45°Rx Feed

Network

Tx Path:

Rx Path:

Cal Path:

Time Shared:

Switch Configuration is Tx

Mode

Power Network

Control Network Control Logic

LPS 1

LPS 2LPS 3

LPS 4LPS 5

Phase Shifter 1

Phase Shifter 3

Phase Shifter 4

90° Hybrid Coupler 2

HPS 2

HPS 1

HPS 3

HPS 4

DA 2

DA 4

HPA 1

HPA 2

HG LNA 1LNA 1

Digital

Attenuator 2

Digital

Attenuator 1

Phase Shifter 2

Phase Shifter 5

DA 1

DA 3

Voltage

Regulation

77dB

Attenuator

77dB

Attenuator

λ/4

λ/4

λ/4

λ/4

HG LNA 2LNA 2

90° Hybrid Coupler 1

LPS = Low Power Switch

DA = Driver Amplifier

HPA = High Power Amplifier

HPS = High Power Switch

HG LNA = High Gain Low Noise Amplifier

LNA = Low Noise Amplifier

Z0

+45° Antenna

Feed

-45° Antenna

Feed

copper flood spacing is more than twice the substrate height to ensure coplanar propagation is not dominant.

14 independent supply voltages are provided for good isolation between components, and to cater for the diversity of voltage supplies (which is a natural result of using COTS components from various manufacturers). A high current variable linear voltage regulator was used for all positive supplies, and a -5V low noise linear voltage regulator was used for the two negative supplies. With this distribution system, 34.56% efficiency in transmit mode at rated output power and 2.66W receive mode power consumption is anticipated.

Thermal management is a critical issue for the module, as both power amplifiers will be operating at rated power. Vias are installed under and around critical components from the RF Layer to the Thermal/Control Layer where heatsinks are mounted with thermally conductive (but electrically non-conductive) paste, as per manufacturer specification. PCB manufacturing issues meant the via density recommended by the manufacturer could not be implemented for several components, but thermal distribution was estimated to be sufficient. However, thermal circuit simulation indicates that in calibration mode the junction temperature of the HPAs is only 0.28°C below maximum, showing that the system should not be put in calibration mode for an extended period of time.

The transmit amplifier chain (without the hybrid couplers) has the structure shown in Fig. 5.

TQP7M9102

2 AP561 1CHP3010-QFG 2

TQP7M9102

1

50Ω50Ω50Ω 50Ω

Figure 5. Amplifier Chain with Known Approximate Impedances

The known impedances are based on manufacturers’ datasheets and signal source and antenna specifications. All amplifiers require a stabilising resistor on the input to produce an unconditionally stable amplifier block to simplify design. Bilateral design via operating power gain and available power gain design methods is performed to achieve the desired gains for each amplifier. Matching networks are implemented using a single capacitor and single inductor as lumped components in an L-section network. C0G Capacitors from AVX and multilayer chip inductors from Toko are used. Series capacitor L-sections are selected where possible to minimise the additional number of DC block capacitors required. Schematic simulations of the combined transmit chain show linear phase as required, and gain shown in Fig. 6.

1215 1235 1255 1275 12951300

Frequency (MHz)

Combined Amplifier Chain Gain

40.5

41

41.5

42

42.5

DB(|S(2,1)|)Combined Amplifier Chain

Figure 6. Combined Transmit Amplifier Chain Gain

As the simulation includes the 7dB insertion loss of the phase shifters, the gain is required to be above 41.155dB (49dB - 7dB - 0.845dB). This design meets the specification for frequencies less than 1.290GHz. This is acceptable because the calibration path will allow the gain deviation to be measured and compensated for in operation.

The control FPGA, in addition to supporting direct control of all bits, also has a Moore state machine programmed to perform a real-time Transmit/Receive operation sequence. This mode provides safety timing to ensure the HPAs do not damage the LNAs when switching from transmit to receive.

The designed, manufactured, and assembled module is shown in Fig. 7.

Figure 7. Prototype Module

III. RESULTS

A. RF Tests

RF tests were performed with a Rhode and Schwarz ZVL3 Vector Network Analyser (VNA) by measuring the S-parameters of RF components and subsystems. A 20dB 25W bi-directional attenuator is used to protect the VNA when measuring high power signals. A 50Ω 15W load is used to load unmeasured paths. The 50Ω transmission lines, digital phase shifters, digital attenuators, LNAs and digital switches functioned as designed and within manufacturer’s specifications. Measurement of the transmit path gain shown in Fig. 8 indicates that the prototype has deviated significantly from the design.

Figure 8. Transmit Path Gain

These measurements show that 15dB-20dB of gain has been lost across the band of interest, and the designed gain peak has shifted down 300MHz. In addition, there is a 5dB gain variation over frequency in the band of interest; in the design it is less than 2dB. The cause of this shift is the reflection coefficients of the matching networks are not as designed; this is due to additional inductance and capacitance in the components themselves, as well as inductance and capacitance introduced by the PCB structure the components are mounted on. These were not taken into account in the design of the matching networks as there was not enough time for EM simulations of the PCB. While the output power does not meet requirements, it is sufficient to characterise the polarisation control mechanism, and ~1W output will be enough to conduct antenna radiation pattern experiments in the future.

B. Polarisation Control Tests

The PCB implementation of the polarisation control circuit demands a revision to the transfer function derived in equation (2). Due to PCB space restrictions, some of the paths assumed identical have different lengths. This will introduce a frequency dependent phase difference between the two paths. The lower path between Hybrid Coupler 1 and Hybrid Coupler 2 is longer than the upper, this phase difference is ϕ1. The distance from Hybrid Coupler 2 to the +45° Antenna port is longer than that to the -45° port, this phase difference is ϕ2. Equation (2) is updated to produce equation (3) by incorporating these phase differences, and converting to S parameters by removing the reference signal voltage term R.

2cos

2sin

1

1

2

4521

4521

2

1

i

i

pred

prede

AeS

S (3)

At 1.215GHz, ϕ1 is predicted to be 78.815° and ϕ2 is predicted to be 113.907°. S21 parameter measurements were made of the transmit path for all possible settings of (α – β) for several frequencies. This measured S21 parameter is compared to the S21 parameter predicted by equation (3), initially setting ϕ1 = ϕ2 = 0. Note that the A term in equation (3) is taken from the actual gain measurements at the respective frequency.

Figure 9. Polarisation Control Measurement Comparison without Offsets

The measured magnitude plot in Fig. 9 matches well in these results. The measured maximum gain is less than the predicted value because the predicted value does not take into account losses in Hybrid Coupler 2 and the HPSs due to test access positioning. The phases have the same general shape, but there are discrepancies. There are also phase deviations when the magnitude becomes small; this is most likely a measurement error in the VNA as the phase accuracy degrades with signal strength. ϕ1 is estimated by calculating the sum of differences squared between the predicted and measured magnitude responses, and estimating ϕ1 to minimise this metric. ϕ2 is found by calculating the sum of differences squared between the predicted and measured phase difference between the paths, and estimating ϕ2 to minimise this metric. Having found ϕ1 = 68.062° and ϕ2 = 119.188° at 1.215GHz

(close to predicted values), the magnitude and phase responses closely correspond, as shown in Fig. 10.

Figure 10. Polarisation Control Measurement Comparison with Offsets

The results of the polarisation control tests validate equation (3), and demonstrate the RF circuit elements are operating as designed.

IV. CONCLUSION

A functional L-band T/R module is successfully implemented. The module can phase shift and amplify signals in receive and transmit, control the gain in receive, and has a flexible switch topology that allows simultaneous H and V reception and access to the transmit/receive loop for calibration

purposes. The module is differentiated from those described in literature by incorporating a novel 360° polarisation control circuit. This circuit determines the polarisation angle by controlling relative phase while keeping HPA output constant. It is demonstrated through measurement that equation (2) correctly describes the output signals of this control system and it was shown that an extra phase shifting element is required to control the phase center of the module. Testing showed the module satisfied most of the Garada project requirements; due to additional impedances of the PCB structure and passive components the transmit path gain was not satisfactory. Despite this, the module is functional and can be used in future projects that could explore multi-module operation and more sophisticated control and calibration systems. The implementation also shows that a COTS approach is a viable strategy for radar prototyping and development.

ACKNOWLEDGMENT

This work was supported by the UNSW Garada Project, Astrium EADS and General Dynamics New Zealand.

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