Post on 02-May-2023
Composite Corporate Traveling-Wave PowerDividers for Slot Array Waveguide Antennas
Dominic Deslandes,1 Francois Boone,2 Ke Wu3
1 Department of Computer Science, Universite du Quebec a Montreal, Montreal, QC, Canada2 Department of Electrical and Computer Engineering, Universite de Sherbrooke,Sherbrooke, QC, Canada3 Department of Electrical Engineering, Ecole Polytechnique de Montreal, Montreal, QC, Canada
Received 24 January 2008; accepted 26 April 2008
ABSTRACT: A composite corporate traveling-wave power divider is presented. The single-
layer structure is composed of three parts: two interdigital traveling-wave subsections com-
bined with a power splitter. An iterative design technique is described in which the divider
is split into a number of basic blocks. Large-scaled networks are then easily designed
because the whole structure does not need to be simulated. A method to take into account
the insertion losses is also proposed and bandwidth enhancement is discussed, which is done
by increasing the number of corporate layers. Experimental results are also shown for a 1:4
subsection. It provides equal output power with 0.5 dB of insertion loss. The phase-shift
between output ports is close to the specifications of 21508 at 30 GHz, with an error of lessthan 28. It is also shown that this topology is well suited for frequency scanning antenna. VVC 2008
Wiley Periodicals, Inc. Int J RF and Microwave CAE 19: 177–186, 2009.
Keywords: antenna feeds; power dividers; rectangular waveguides; slot arrays; traveling-wave
devices
I. INTRODUCTION
Slot array waveguide antennas are widely used in
communication and radar systems. These antennas
are composed of two parts, namely, an array of slots
and a feed network. Many different topologies of
feeders have been proposed over the last years. The
most common technique to distribute the power is to
use the well-known corporate feeder. However, in
high-gain antenna design, a large number of parallel
waveguides must be used and the feeders become
bulky. A more compact feed network can be obtained
with a perpendicular feeder located under the antenna
waveguides. The coupling is then achieved with the
help of alternatively inclined slots to obtain an in-
phase excitation [1]. This structure is very compact
but requires the precision manufacturing and align-
ment of two pieces.
A similar feeder realized on a single-layer was
also proposed [2]. The perpendicular feed waveguide
is then coupled to the waveguide array with irises.
This structure is, however, limited to alternating-
phase fed antennas, thus preventing the design of a
broadside array. This problem can be circumvented
by coupling two waveguides together instead of one,
as shown in [3]. The coupling irises are then spaced
one wavelength apart and all the waveguides of the
array are fed in-phase. But, it is impossible to design
a feeder with different phase shifts between the out-
puts in this configuration. An inclined feeder can pro-
vide certain flexibility in the choice on the output
Correspondence to: D. Deslandes; e-mail: deslandes.dominic@uqam.ca
DOI 10.1002/mmce.20338Published online 20 August 2008 in Wiley InterScience (www.
interscience.wiley.com).
VVC 2008 Wiley Periodicals, Inc.
177
phases [4]. Nevertheless, this last structure is bulky
and does not allow designing feeders with an arbi-
trary combination of phase and amplitude.
On the other hand, traveling-wave structure has
been successfully applied to design compact power
divider/combiner for high-power amplifier applica-
tions [5]. For these applications, the relative output
phase shifts of the divider are not important since it is
compensated by the combiner. The outputs can then
be spaced apart which decreases the effect of higher-
order modes. However, for antenna applications, the
antenna waveguides must be placed closely to avoid
grating lobes. The design then becomes much more
complex due to the higher-order mode effects. These
effects may be circumvented by using an interdigital
structure, as applied in the design of dual-polarization
antennas [6].
The topology presented in this article is not
purely traveling-wave but a composite corporate
traveling-wave structure. A topology using this
scheme has been proposed and was briefly dis-
cussed by the authors in [7]. This article is to extend
the work presented at the conference by including
the evaluation of the coupling level, a method to
take into account the insertion losses, a comprehen-
sive description of the design steps, and a discus-
sion on bandwidth enhancement. The topology and
operating mechanism are discussed in Section II.
The iterative design process is detailed in Section
III. Section IV deals with the concept of bandwidth
enhancement and an experimental validation is pre-
sented in Section V.
II. POWER DIVIDER TOPOLOGYAND MECHANISMS
An exploded view of the composite corporate travel-
ing-wave power divider topology, including the slot
array waveguides, is shown in Figure 1. The power
divider topology can be divided in three parts: two
interdigital subsections joined by a power splitter, all
identified in Figure 1 by numbers 1–3, respectively.
The two interdigital subsections are composed by
a cascade of T-junctions interconnected by wave-
guide sections. In Ref. [2], the length between two T-
junctions was around half a wavelength. The interac-
tion of higher-order modes is then non-negligible and
the structure must be optimized in a single bloc, sig-
nificantly increasing the simulation time. The struc-
ture of Figure 1 has solved this problem by using an
interdigital topology. The spacing between two con-
secutive T-junctions of a subsection is then approxi-
mately a wavelength. The higher-order modes are
sufficiently attenuated to be neglected and the feeder
can be decomposed in N separated T-junctions joined
by N 2 2 straight waveguides, where N is the number
of antenna waveguides in the array. The problem
then resumes to the design of N T-junctions which,
when cascaded, will provide the right output ampli-
tudes and phases.
In Ref. [7] it has been proved that the proposed to-
pology is universal, which means the structure can be
used to design feeders with any arbitrary combina-
tions of amplitudes and phases. This property is
obtained by using two different mechanisms. First,
the T-junctions are set to provide the right amplitude
level and secondly, widths of waveguide sections
interconnecting T-junctions are adjusted to fix the
phase shift. Thus, each block of the two interdigital
sub-sections is designed to adjust only one parameter
(amplitude or phase). This leads to the use of an itera-
tive design technique to fix the coupling levels and
connecting waveguide widths, which will be pre-
sented in Section III.
Figure 1. Exploded view of the composite corporate
traveling-wave power divider including the slot array
waveguide antenna: (1) first interdigital sub-section; (2)
second interdigital sub-section; (3) power splitter.
178 Deslandes, Boone, and Wu
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
The last part of the divider is the power splitter
subsection. This is realized with a power splitter and
two bends. There are many different ways to achieve
these functions. First of all, the power splitter can be
realized with a full-height post [8], a partial-height
post [9], a stub [10], or different matching structures
used to obtain the right output levels. The corner is
realized with a curved bend [11], a mitered bend
[12], or with posts [13]. Different combinations are
shown in Figure 2.
It is important to realize that this feeding subsec-
tion must not always provide an in-phase division. In
fact, the required phase is related to the number of
slots in each waveguide. Figure 3 shows two consec-
utive antenna waveguides in the array. The arrows
over the slots represent the polarization of the electric
field in the slots. The phase of the electric field
required to adequately excite the slots is shown
beside each slot, in the center of the antenna wave-
guides. This phase is either 08 or 1808. When the
number of slots is odd, all antenna waveguides must
be excited in-phase as described in Figure 3a. The
power splitter must then provide an in-phase division.
However, if the number of slots is even, the two
interdigital subsections must be excited out of phase,
as indicated in Figure 3b. This is easily realized by
offsetting the power-splitter by a quarter wavelength,
as shown in Figure 4.
III. ITERATIVE DESIGN TECHNIQUE
Different techniques have been used to synthesize
slot array waveguide antenna feeders. Coupling-slot
feeders have been analyzed with the method of
moments and a pattern synthesis technique was used
Figure 2. Three different topologies of power splitter:
(a) mitered bends with stepped power-splitter; (b) rounded
bends and splitter; (c) bends and splitter realized with
posts.
Figure 3. Required phase at the input of antenna waveguides: a) if the number of slots is odd,
waveguides are fed in-phase; b) if the number of slots is even, the waveguides are fed out of phase.
Corporate Traveling-Wave Power Dividers 179
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
to obtain a shaped beam broadside antenna [1]. Pla-
nar waveguide feeders have been analyzed with nu-
merical technique including method of moment [2, 3]
and finite element method [4]. The synthesis is
achieved by successively designing and cascading
each junction [2–4]. An iterative procedure can then
be applied to improve the results [2]. This last tech-
nique is suitable for traveling-wave structures
because each junction parameter is dependant of all
the others junctions.
In the proposed divider, each interdigital subsec-
tion is decomposed into basic building blocks, as
shown in Figure 5a. Since the length of the inter-
connecting waveguides is about a wavelength, a
single mode representation is sufficient and the
analysis of the whole structure can be done by cas-
cading 2 3 2 S-matrix, as illustrated in Figure 5b.
However, all building blocks are not independent
and an iterative technique must be applied to take
into account mutual coupling effects. It is important
to note that the numbering of T-junctions begins at
the short-circuit side and not at the input port side
as shown in Figure 5a.
A. Evaluation of the Coupling Level
The T-junction shown in Figure 6 can couple any
power level between 0 dB and 21 dB. In the spe-
cific case of 0 dB, the Port 2 is short-circuited and,
with a proper design, all the power is transferred to
Port 3. This is always the case for junction number 1.
When the coupling level tends to be 21, the iris
width and inset length tend to be zero, which gives a
straight waveguide section. In all other cases, the iris
width is used to adjust the coupling level and the
inset length and offset are used to match the junction.
The first step in the design process is to evaluate
the required coupling level in every T-junction. This
level is fixed by the power distribution required to
obtain the desired radiation pattern. This distribution
can be uniform, binomial, Chebychev, etc. In all T-
junctions, the incident power at Port 1 is completely
transferred to Ports 2 and 3, if the losses are neglected.
Since all the junctions are perfectly matched at Port 1,
there is no reflected power at Port 1 and, consequently,
no incident power at Port 2. This comes from the fact
that all the junctions are cascaded and Port 2 of junc-
tion number n þ 1 is connected to Port 1 of junction
number n. Also, there is no incident power at Port 3 at
the central frequency since it is connected to a
matched antenna. In all T-junctions, the transmission
coefficient between Ports 1 and 3, S3l, n, can be written
in the form of a power ratio:
S3l;n�� ��2¼ P3;n
Pl;n; ð1Þ
where Pl,n is the incident power at Port 1 and P3,n is thetransmitted power at Port 3, both for the nth junction. Fur-thermore, the input power of the nth junction is equal to
the sum of the output power of the preceding junctions:
Figure 4. Phase adjustment for the power splitter: (a) in-
phase outputs for an odd number of slots; (b) out-of-phase
outputs for an even number of slots.
Figure 5. Decomposition of the power divider into basic blocs: (a) physical representation; (b)
electrical representation.
180 Deslandes, Boone, and Wu
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
Pl;n ¼Xnk¼1
P3;k� ð2Þ
Combining (1) and (2), we obtain:
S3l;n�� ��2¼ P3;nPn
k¼1P3;k
: ð3Þ
However, the amplitude distribution of feeder net-
works is more often given in terms of voltage other
than power. In this case (3), becomes equal to:
S3l;n�� ��2¼ V2
nPnk¼1
V2k
; ð4Þ
where Vk is the desired voltage for the array
shaping.
B. Insertion Losses
When there are a large number of outputs in the
power divider, the insertion loss can yield influence
on the amplitude ratio. The traveling wave will be
more and more attenuated as it propagates through
the divider. It is, however, possible to take it into
account in the design process. Equation (1) can be
rewritten in the following form:
P3;n ¼ Pl;n S3l;n�� ��2: ð5Þ
Also, the output power of the first T-junction can be
expressed as a function of the input power of the nth
junction:
P3;l ¼ Pl;n S3l;l�� ��2Yn
k¼2S2l;k�� ��2 S2l;ðk;k�1Þ�� ��2: ð6Þ
In the last equation, S3l,l is the transmission coeffi-
cient between Ports 1 and 3 for the first T-junction,
S2l,k is the transmission coefficient between Ports 1
and 2 for the kth T-junction and S2l,(k,k2l) is the trans-
mission coefficient of the connecting waveguide
between the kth and k21th T-junctions. Dividing (6)
by (5) and rearranging it, we obtain:
S3l;n�� ��2¼ P3;n
P3;lS3l;l�� ��2Yn
k¼2S2l;k�� ��2 S2l;ðk;k�1Þ�� ��2: ð7Þ
If a voltage distribution is used in place of the power
distribution, we obtain:
S3l;n�� ��2¼ V2
n
V2l
S3l;l�� ��2:Yn
k¼2S2l;k�� ��2 S2l;ðk;k�lÞ�� ��2: ð8Þ
C. Iterative Design Technique
The relative phase shift between two consecutive T-
junctions in an interdigital subsection, by neglecting
the internal phase shift of the T-junction, is given by:
D/� 2mp ¼ �bal0; ð9Þ
where ba is the propagation constant of the feeder
waveguide, l0 is the spacing between two adjacent T-
junctions of a subsection and m is a constant that has
Figure 6. T-junction basic building block: (a) ports and
reference plane; (b) dimensions.
Corporate Traveling-Wave Power Dividers 181
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
been defined in [7]. The propagation constant is equal
to:
ba ¼ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffik20 �
pa
� �2:
rð10Þ
The distance between two T-junctions can be found
as:
l0 ¼ 2bþ 2g; ð11Þ
where b is the antenna waveguide width and g is the
thickness of walls between these waveguides.
The phase shift between Ports 1 and 2 as well as
Ports 1 and 3 (ffS2l and ffS3l), referring to Figure 6a,
varies with the coupling value of the T-junction.
They must then be taken into account with an itera-
tive technique. First of all, variables cn, dn, and en areoptimized for every T-junction under the following
conditions:
1Þ Minimize S1l;n�� ��
2Þ S3l;n ¼ 10 logV2nPn
k¼1V2k
0BB@
1CCA:
ð12Þ
If the insertion losses are not negligible, the follow-
ing equation must then be used in place of (12):
1Þ Minimize S1l;n�� ��
2Þ S3l;n ¼ 10 logVnVl
S3l;l�� ��2:Yn
k¼2S2l;k�� �� S2l;ðk;k�lÞ�� �� !
:
ð13Þ
The values of ffS2l,n and ffS3l,n calculated in this first
iteration, with the reference plane as defined in Fig-
ure 6a, are then applied to resize the width of wave-
guide sections ain,nþ1, as follows. The length of the
waveguide section between T-junction numbers nand n þ 1, at the iteration step i, is given by:
lin ¼ l0 þ ein � einþ1; ð14Þ
where l0 is defined in (11) and ein is the offset length
of inset number n at iteration number i. When the in-
ternal phase shifts of T-junctions are taken into
account, (9) becomes:
D/n;nþ1 � 2mp ¼ ffSi2l;nþ1 þ ffSi3l;n � ffSi3l;nþ1 � binlin:ð15Þ
Solving (15) for bin and combining with (10), the
width of interconnecting waveguide section number nfor the next iteration can be found by
aiþ1n;nþ1 ¼pffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi
k20 � bin� �2q : ð16Þ
The widths of ports 1 and 2 in every T-junction are
then set to be their new values for the next iteration.
D. Design Steps
The design steps required to obtain the power divider
with specifications are:
1. Estimate the initial dimensions: At the first
step, variables a, b, and g must be selected in
order to respect (9) combined with (10) and
(11). In other words, the waveguide dimen-
sions and spacing must be selected to obtain
the specified phase shift when the T-junction
internal phase shifts are neglected. There are
indeed multiple solutions and the designer
must take into account other parameters in
his/her choice, as mechanical feasibility, gra-
ting lobes, single mode propagation, physical
interconnection, etc.
2. Initialize all variables: Then, the initial
dimensions, referring to Figure 6, are set to be
cn 5 an, n þ 1/2, dn 5 an, n þ 1/2, and en 5 0.
3. Evaluate the required coupling level values:The coupling levels are evaluated by using (4)
or (8) if insertion losses are not negligible.
For the later, the values of S2l,k are not
known, so they are preliminarily set to be
zero. However, it is possible to initially evalu-
ate the losses in the waveguide sections inter-
connecting the T-junction, S2l,(k, k þ 1). Wave-
guide width an and length l0 have been esti-
mated in step 1 and the insertion losses is
readily evaluated [14].
4. Optimize all T-junctions: All the T-junctions
are optimized by following the criteria defined
in (12) or (13) if the insertion losses are not
negligible. Only parameters cn, dn, and en are
optimized. For the first T-junction, however,
the length of the short-circuited stub placed at
Port 2 must also be optimized.
5. Compute the width of the waveguide sections:The width of the waveguide section is reeval-
uated, taking into account the internal phase
shift through (15).
182 Deslandes, Boone, and Wu
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
6. Verify the convergence: The error functions
for the kth section, evaluated between two con-
secutive iterations, are given by:
Ek ¼ aiþ1k � aik�� ��: ð17Þ
To obtain the convergence, the error for all wave-
guide sections must be smaller than a defined toler-
ance which should also be smaller than the manufac-
turing tolerance. The flowchart of the design process
is depicted in Figure 7.
IV. BANDWIDTH ENHANCEMENT
The design process shown in Figure 7 is achieved
only at the central frequency. Therefore it cannot
assess accurately the bandwidth of the structure. This
bandwidth is limited by the frequency variation of
the matching and coupling levels of T-junctions.
Interestingly, it has been found that the T-junction to-
pology shown in Figure 6 almost provides a constant
coupling level over the entire waveguide bandwidth.
Figure 8 shows the frequency response of a T-
junction optimized to provide a 3 dB coupling at
30 GHz. We can see that the bandwidth is not limited
by the variation of the coupling level but by the
matching level. This one is below 220 dB from 27.4
GHz to 32.8 GHz. Similar behaviors have been
observed for other coupling values.
The topologies shown in Figure 5a can be used to
design antenna feeders with any number of outputs.
However, since this is a composite corporate travel-
ing-wave feeder, the output phase will vary according
to frequency. This phase shift will tilt the main beam
angle. When a large number of outputs are required
or a wide bandwidth is necessary, the composite cor-
porate traveling-wave feeder bandwidth can be
improved by increasing the number of layers of the
corporate part. This concept is illustrated in Figure 9
with a 1:8 power divider.
In Figure 9a the corporate part has one layer and
there are two interdigital sub-sections. In Figure 9b
the corporate part has two layers and there are four
interdigital sub-sections. In Figure 9c the feeder is
completely corporate. In the later case, the bandwidth
is no longer limited by the output phases as they are
all fed in-phase over the entire frequency span. In
this way, it is easier to obtain the best compromise
between size and bandwidth.
Figure 8. Frequency response of a T-junction optimized
to provide 3 dB coupling at 30 GHz.
Figure 7. Flow-chart of the iterative design process.
Corporate Traveling-Wave Power Dividers 183
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
V. EXPERIMENTAL VALIDATION
The proposed technique has been validated on a 1:8
power divider working at 30 GHz with a uniform am-
plitude distribution and a phase shift of 2758between adjacent antenna waveguides. The feeder is
split in three blocks: two identical interdigital 1:4
subsections linked together with a 1:2 power splitter.
The problem is then simplified to the design of a 1:4
power divider with a uniform amplitude distribution
and a phase shift between adjacent outputs of 21508,that is to say D/n,nþ1 5 22.62 rad for all n. Bycombining (9) and (10) with a 5 7.112 mm (WR28),
f 5 30 GHz, and D/ 5 22.62 2 2p 5 28.9 rad, one
gets l0 5 19.894 mm. A possible combination to
solve (11) is b 5 7.112 mm and g 5 2.835 mm. For
a uniform distribution, Vn 5 1, the coupling level is
found by using (4) which gives these couplings: 0
dB, 23 dB, 24.77 dB, and 26 dB. The design proc-
ess is then applied and the dimensions for all itera-
tions have been reported in [7]. Five iterations are
required to reach the convergence for a variation
lower than 0.1% for all dimensions. Even though the
convergence is not guaranteed, we did not observe
any divergence problems in the different designs we
tested. We also observed that the required number of
iterations is proportional to the number of outputs.
An interdigital subsection has been fabricated and
measured to validate the proposed structure as wellas the design technique. The prototyped structure isshown in Figure 10 without the cover. The measure-ment of this 5 port network is performed with three
waveguide loads and two coaxial transitions. Theposition of the transitions and loads are then per-muted to measure all the reflection and transmissioncoefficients. The main disadvantage of this technique
is that results are highly dependant of the return lossof loads. The measured coefficients of reflection forthe three loads in the bandwidth of interest are shown
Figure 10. Picture of the fabricated traveling-wave sub-section.
Figure 9. Bandwidth enhancement by increasing the
number of corporate layers: (a) one corporate layer; (b)
two corporate layers; (c) completely corporate.
184 Deslandes, Boone, and Wu
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
in Figure 11. The return loss is very good but there
are strong ripples in the entire measured bandwidth.
These ripples will slightly affect the measured results,
as shown in the next two figures.
The measured results are shown in Figure 12. The
return loss is better than220 dB from 26.9 GHz to 31.2
GHz. All transmission coefficients fluctuate around
26.5 dB, which represents 0.5 dB insertion loss. The
phase shifts between adjacent outputs are shown in Fig-
ure 13. The structure provides a phase shift around
21508 at 30 GHz, with a maximal error of 28.A characteristic behavior of this topology is the fre-
quency dependence of the phase shift between the out-
puts. As described in Section IV, this aspect can reduce
significantly the bandwidth. However, it can also be
used to design frequency scanning antennas. In this
case, the frequency dependence is desired and should be
linear. Figure 13 shows that the subsection provides a
3608 shift from 26 GHz to 36 GHz with good linearity.
VI. CONCLUSION
A novel composite corporate traveling-wave power di-
vider has been proposed and presented in this article.
The modular structure can be designed to provide any
combination of phases and amplitudes. With the itera-
tive design technique proposed in this work, the feeder
is divided into several blocks. Each block is then ana-
lyzed and optimized separately at only one frequency.
In this way, the complete structure would never have
to be simulated, which speeds up the design process.
Furthermore, a technique to overcome the loss
effects on the output amplitudes has been developed.
It opens the way to the design of large-scaled but
compact antenna feeders, without distortion due to
conductor losses in the traveling-wave section.
Experimental results have validated the proposed
structure topology as well as the developed design
process. With the increasing popularity of millimeter-
wave imaging, this structure provides a compact solu-
tion for the design of frequency scanning antennas.
ACKNOWLEDGMENTS
The authors wish to acknowledge the financial support of
the Natural Sciences and Engineering Research Council of
Canada.
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BIOGRAPHIES
Dominic Deslandes received the B.Sc.
degree in electrical engineering from the
University of Sherbrooke, Canada, in
1998, and the M.Sc. and Ph.D. degrees
from the Ecole Polytechnique de Mon-
treal, Canada, in 2001 and 2005, all in
electrical engineering. From 2006 to
2007, he was a Post-Doctoral Researcher
at the University of Sherbrooke, Canada.
In December 2007, he joined the Department of Computer Sci-
ence at the University of Quebec in Montreal, Canada, where he
is now assistant professor. His research interests include the anal-
ysis, synthesis, and integration of passive and active components
for microwave and millimeter-wave systems. Dr. Deslandes was
awarded the Natural Sciences and Engineering Research Council
Doctoral Prize (Best Engineering Thesis in Canada) in 2007 and
the Best Student Paper Award at APMC in 2005.
Francois Boone received the Eng. Dipl.
in Electrical Engineering from Ecole
Nationale Superieure d’Electronique,
d’Electrotechnique, d’Informatique et
d’Hydraulique de Toulouse (ENSEEIHT),
France, in June 1992 and the DEA in
microwave engineering from the Institut
National Polytechnique de Toulouse
(INPT) in 1993. In 1997 and 2000,
respectively, he received M.Sc. and Ph.D degrees in microwave en-
gineering, both from Ecole Polytechnique de Montreal, Canada in
October 1999, he joined the Department of Electrical and Computer
Engineering of Faculty of Engineering at the University of Sher-
brooke, Canada, where he is now an associate professor. His research
interests concern analysis, modeling and design of passive and active
microwave, and millimeter-wave components and circuits.
Ke Wu is Canada Research Chair Profes-
sor in RF and millimeter-wave engineer-
ing at Ecole Polytechnique (University of
Montreal). He has been Director of the
Poly-Grames Research Center. He has
authored or coauthored more than 600
referred papers, and a number of books/
book chapters and patents. He has held
key positions in and has served on various
panels and international committees including the chair of many
committees and international conferences/symposia. He has
served on the editorial/review boards of many journals, transac-
tions, and letters including associate editor, editor, and guest edi-
tor. He is an elected IEEE MTT-S AdCom member for 2006–
2009 and serves as chair of the IEEE MTT-S Transnational Com-
mittee. Dr. Wu was the recipient of many awards and prizes. He
is a Fellow of the IEEE, a Fellow of the Canadian Academy of
Engineering and a Fellow of the Royal Society of Canada (The
Canadian Academy of the Sciences and Humanities).
186 Deslandes, Boone, and Wu
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce