New Wideband Miniature Branchline Coupler on IPD Technology for Beamforming Applications

IEEE TRANSACTIONS ON COMPONENTS, PACKAGING AND MANUFACTURING TECHNOLOGY, VOL. 4, NO. 5, MAY 2014 911

New Wideband Miniature Branchline Coupler onIPD Technology for Beamforming Applications

Diane Titz, Member, IEEE, Fabien Ferrero, Member, IEEE, Romain Pilard, Member, IEEE, Claire Laporte,Sébastien Jan, Hilal Ezzeddine, Frédéric Gianesello, Member, IEEE, Daniel Gloria,

Gilles Jacquemod, Member, IEEE, and Cyril Luxey, Senior Member, IEEE

Abstract— In this paper, we present a new wideband minia-ture branchline coupler as a key circuit to be integrated in60-GHz packaged beamforming networks for phased-arrayantennas. First, the integrated passive device (IPD) technologyfrom STMicroelectronics is investigated in the mm-wave rangethrough the simulation, fabrication, and measurements of amicrostrip line and a simple hybrid coupler. Then, a novel couplertopology with emphasis on miniaturization and broadband opera-tion is theorized. Analytical equations are derived and a 60-GHzcoupler is optimized on IPD technology. Measurement resultsare discussed and compared with state-of-the art publications.The whole 57–66-GHz bandwidth is efficiently covered withthe three following performance: −10-dB impedance matching,±1-dB amplitude imbalance, and ±5° phase imbalance. As anapplication example, the novel coupler is integrated into a 4 × 4Butler matrix suitable for an array-antenna demonstrating state-of-the art performance in terms of insertion loss and phase error.The measurement of different samples shows low variation ofthe IPD process because of very good reproducibility making ita suitable candidate for circuits operating in the 60-GHz band.

Index Terms— 60 GHz, array antenna, beamforming network,Butler matrix, coupler, IPD technology.

I. INTRODUCTION

AN INCREASE interest in the 60-GHz band (57–66 GHz)has emerged for high-data-rate communications. Conse-

quently, several standards have been issued [1] and siliconBiCMOS transceivers development have shown that suitableperformance is indeed possible. However, some issues are stillto be addressed concerning the cointegration and packaging

Manuscript received April 28, 2013; revised December 30, 2013 andFebruary 24, 2014; accepted February 25, 2014. Date of publication March 27,2014; date of current version May 1, 2014. This work was supported in partby the CIMPACA Design Platform and in part by CREMANT. Recommendedfor publication by Associate Editor D. G. Kam upon evaluation of reviewers’comments.

D. Titz and G. Jacquemod are with EpOC, Université Nice SophiaAntipolis, Sophia Antipolis 06903, France (e-mail: [email protected];[email protected]).

F. Ferrero is with the Centre de Recherche Mutualisé sur les AntennesLaboratoire Commun entre l’Université Nice Sophia Antipolis, le CNRS etOrange Labs La Turbie, Université Nice Sophia Antipolis, Valbonne 06560,France (e-mail: [email protected]).

R. Pilard, S. Jan, F. Gianesello, and D. Gloria are with STMicroelectronics,Crolles 38926, France (e-mail: [email protected]; [email protected];[email protected]; [email protected]).

C. Laporte and H. Ezzeddine are with STMicroelectronics, Tours 37071,France (e-mail: [email protected]; [email protected]).

C. Luxey is with EpOC, Université Nice Sophia Antipolis, Sophia Antipolis06903, France, and also with the Institut Universitaire de France, Paris 75005,France (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TCPMT.2014.2311092

of the integrated circuits (ICs), the passive circuits, andantennas [2].

This paper focuses on the design and the integration of abroadband passive circuit at mm-wave frequencies: the hybridcoupler. Due to the short wavelength at 60 GHz (λ0 = 5 mm),using quarter-wave length lines is not prohibitive in termsof occupied space. These couplers are often used in mixers,amplifiers, vector modulators, and are also very useful inbeamforming networks where electronically scanned antennaarrays might overcome the inherent 60-GHz path loss andalleviate nonline-of-site communications. Among the bestavailable solutions, reflection-type phase shifters [3], [4] andButler matrices [5] are very often employed. In those networks,the necessary couplers must be small and wideband enoughto both maintain satisfactory system performance as well as ahigh level of integration. The technological choice is also tobe discussed. When using passive circuits, the insertion lossis often the limiting factor. Silicon substrate implementationis a possible option but still has some drawbacks [6]–[9].The integrated passive device (IPD) technology proposed inthis paper is used for low-loss and low-cost integration ofminiaturized couplers [10]. However, in the mm-wave regime,it has being mainly characterized for antennas [11]–[14], filters[15], [16], and an antenna integrated within a filter [17]. So far,from the best of the authors’ knowledge, one IPD couplerdesign was recently published in X-band [18] and only tworecent attempts have been made in V-band [19], [20].

In this paper, we present a new wideband miniature branch-line coupler (BLC) operating at 60 GHz in IPD technologyfrom STMicroelectronics. First, a technological study of theIPD process for mm-wave communications is proposed. Then,theory and equations of the novel coupler are presented.A design at 60 GHz is described, measured and comparedwith state-of-the art recently published couplers. Finally, theintegration of this novel coupler into a Butler matrix forphased-array applications is presented. Measurement resultsand comparison to state-of-the-art recently published Butlermatrices confirm competitive performance for such a technol-ogy at V-band.

II. STUDY OF THE IPD PROCESS AT

MM-WAVE FREQUENCIES

The IPD process developed by STMicroelectronics hasbeen especially created to design lossless passive devices likeresistors, capacitors, or inductors. Design of passive circuitslike baluns, transformers, and filters at lower frequencies are

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912 IEEE TRANSACTIONS ON COMPONENTS, PACKAGING AND MANUFACTURING TECHNOLOGY, VOL. 4, NO. 5, MAY 2014

Fig. 1. IPD process build-up with detailed side-view of the BEoL(not to scale).

already sold on the market. Couplers are passive circuits,which can benefit from the IPD technology. A simplified sideview of the back-end-of-line (BEoL) of this process is shownin Fig. 1.

A. 50-� Microstrip Line in IPD Technology

As the IPD technology characteristics (permittivity andlosses) are only known at low frequencies, we focused abasic structure to be able to characterize the technology athigher frequencies. A 12-μm wide 50-� microstrip line wassimulated with HFSS, realized, and characterized. The signalline was etched on the 3-μm-thick M3 layer and its groundplane on the 6-μm-thick M2 layer. Only M2 and M3 copperlayers were used because the M1 layer is made of aluminumand is thinner, which induces a lower conductivity.

The layout of the line and its measured results comparedwith HFSS simulation are shown in Fig. 2. The measurementwas performed using on-wafer probing and a setup composedof two GSG Infinity probes and a 67-GHz VNA. Between55 and 65 GHz, the simulation and measurement curves fitperfectly each other. Therefore, we could conclude that thepermittivity of the benzocyclobutene (BCB) is close to 2.7and its loss tangent lower than 10−3. In turn, the line exhibitsan insertion loss of 0.75 dB/cm, which is very satisfactoryat 60 GHz. Considering the permittivity of the BCB, theguided wavelength was computed to be close to 3.0 mm at60 GHz. Five different samples of the same microstrip lineswere measured showing an excellent reproducibility of theprocess.

B. Branch-Line Coupler

A classical BLC was also designed and fabricated at60 GHz. The layout is shown in Fig. 3. It can be seen thatthe vertical and the horizontal lines of the couplers wereslightly meandered to reduce the occupied area: 563 μm ×574 μm = 0.323 mm2 without the probing pads (instead of0.56 mm2 if the lines have not been meandered). It shouldbe noted that the coupling between the branches was consid-ered during the optimization stages of the coupler through

Fig. 2. (a) Top-view of the layout of the 50 � line. (b) Measured |S11| and|S21| parameters (dots) compared with simulation (thick lines).

Port 1

Port 2

Port 4

Port 3

35Ω line

50Ω line

563 μm

574 μm

Fig. 3. Top-view of the layout of the BLC.

Fig. 4. Simulated S-parameters of the BLC with (solid lines) and without(short dashed line) the signal pads.

the use of a full-wave simulator. The measurement of thecoupler was performed using the same probing setup. M2layer was especially hollowed on purpose, below each signalpads to avoid any parasitic capacitance. Hence, measurementdeembedding was not necessary. To confirm this assumption,HFSS simulations, including the pads, were performed. Fig. 4shows the simulated performance of the coupler with and

TITZ et al.: NEW WIDEBAND MINIATURE BLC ON IPD TECHNOLOGY 913

Fig. 5. Simulated (solid lines) and measured (dotted and dashed lines)reflection coefficient and isolation of the BLC shown in Fig. 3. Measurementsare performed for four different samples.

without the pads. It shows very similar results, demonstratingthat the parasitic capacitance is indeed quite low.

Fig. 5 presents the measurements and simulation of thereflection coefficient and isolation of four realized BLCsamples. It can be seen that the reproducibility of the processis excellent. The performance of one of those samples iscompared to the HFSS simulations in Fig. 6. Despite someripples oscillating around the expected simulated values andeven though the calibration procedure was repeated severaltimes, the agreement is quite fair. The overall performanceof the BLC is summarized in Table I. The insertion loss ofthe BLC is better than 1.4 dB on the 57–66-GHz band. The−10-dB matching bandwidth ranges from 53 to 67 GHz,the ±1-dB amplitude imbalance bandwidth ranges from 56to 65 GHz, and the ±5° phase difference bandwidth rangesfrom 54.5 to 67 GHz.

III. THEORY AND DESIGN OF THE NOVEL COUPLER

The BLC solution is usually avoided when compactness isneeded because of its large size. However, compared with acoupled-line and a Lange coupler, its implementation is lessdependent on the process variation and easier to achieve eitheron multilevel or single metallization processes. To cover the57–66-GHz band, wideband performance is necessary. The useof an all-lumped component topology is then excluded becauseof narrowband characteristics. Our approach was then to takeadvantage of both microstrip lines and lumped capacitors as anefficient mix to achieve a small footprint and wide bandwidthcharacteristics.

A. Equations

The novel coupler topology (Fig. 7) is based on twocascaded quasi-lumped quadrature couplers (QLQC) [21], [22]with a short-circuited stub inserted between those two couplersfor matching and miniaturization purpose. The novel widebandtechnique we focused on is a mix of the following variousdesigns [23]–[25] but we emphasized simplicity.

The novel coupler is made of capacitances (C1 and C2)and transmission lines (TLs) of characteristic impedances andlengths: Za, La, Zb, Lb, and Zc, Lc. At this stage of thesynthesis, we kept the capacitances C1 and C2 at the samevalue, C . The architecture originates from the QLQC topology(Fig. 7, left corner). Analytical equations for the design of a

Fig. 6. Simulated (solid lines) and measured (dotted lines) S-parameters, PI,and amplitude imbalance of one BLC sample from Fig. 3.

QLQC using the even and odd modes decomposition [26] arepresented in [21] and [22]. The coupling coefficient is thengiven by (1) where Ya is the admittance of the TL and Y11is given by (2). For a 3-dB coupler (k = − j ) and for 50 �lines, we obtain Za = 1, La = π /4 (where La is the electricallength in rad), and Y11 = j

k = S21

S31= 1

2

⎛⎜⎜⎝

2Ya(1 + Y11) cos(La)

Y11(Ya cos(La) + j (1 + Y11) sin(La))

+ j (1 + Y 2a + 2Y11(1 + Y11)) sin(La)

Y11(Ya cos(La) + j (1 + Y11) sin(La))

⎞⎟⎟⎠ (1)

Y11 = 2Cω. (2)

From this original design, each time we designed a newcoupler using ADS as the simulation tool, we first set theisolation and reflection coefficients to zero and the outputamplitude imbalance being 0 dB at 60 GHz as the fourgoals to obtain the initial solutions (dimensions) for the TLs.Then, the second step consisted in tuning those values for a+1-dB amplitude imbalance goal (between the outputs) at


TABLE I

SUMMARY OF THE MEASURED PERFORMANCE OF OUR COUPLERS AND STATE-OF-THE-ART COUPLERS ON IPD TECHNOLOGY AT 60 GHZ

Fig. 7. New coupler topology.

Fig. 8. From the QLQC topology to two cascaded QLQCs.

60 GHz. This procedure was found to be efficient to broadenthe operating bandwidth of the coupler.

The first step was to cascade two QLQCs (Fig. 8) to increasethe bandwidth of a single QLQC. If we keep 50 � lines andLa = π/4, the coupling k reduces to the analytical valuein (3), which gives us a value of Y11 = j/2 for a 3-dBcoupling. The wideband behavior of the cascaded QLQC canbe observed in Fig. 9 where it is compared with the behaviorof a single QLQC. The phase imbalance (PI) and reflectioncoefficient behavior are improved but the ±1-dB BW is stillquite narrowband

k = − (−2Y 211 − 2Y11 + jY11 − 1)(−Y11 + j)

Y11(2 j − 2Y11 + 3 jY11 − 2Y 211)

. (3)

A further improvement can be realized using a capacitanceof 2C at the junction between the two QLQCs. However, thissolution might increase the mismatch between the capacitorsand their sensitivity versus the global design. To improve thereliability of the design and add a degree of freedom, we

thought about using a TL between the two central capacitances(Fig. 10). If the characteristic impedance of this line (Zb) isequal to 50 �, then the previous results still apply. If we keepthe values of Za, La , and C of the QLQC, tuning only Zb

and Lb accordingly to (4) gives us a 3-dB coupler behavior.However, the isolated and direct ports are now swapped: Port 2becomes the isolated port and Port 4 the direct port. The bestresults are obtain when L ′

b � 130°, thus Z ′b = 17 �. For

those values, the amplitude imbalance is equal to 0 at 60 GHz.However, as stated before, to maximize the ±1-dB bandwidth,we set L ′

b to 127° and Z ′b to 16 �

Y ′b =

1 +√

1 + sin2(L ′b)

sin(L ′b)

. (4)

The simulation performance of this enhanced coupler iscompared with the performance of a single QLQC in Fig. 11.

However, a microstrip line with these characteristics(L ′

b = 127° and Z ′b = 16 �) would be very large and long and

this coupler would not be miniaturized anymore. One way tosolve this issue is to replace this microstrip line by an equiv-alent combination of a microstrip line ((Zb, Lb)) and a stub((Zc, Lc)), as shown in Fig. 7. We choose a short-circuitedstub for a shorter electrical length in our configuration. Bykeeping the initial values of Za, La, and C of the QLQC, andtuning only Zb, Lb, Zc, and Lc accordingly to (5) gave us asatisfactory 3-dB coupler behavior

− jYc

tan(Lc)= −3Y

′b( jY

′b + (cos(2La) + j sin(2La))

× (Y′b cos(2L

′b) + j

1 + Y′2b

2sin(2L

′b)))/

(cos(La)+ j sin(La)2(Y′b cos(L

′b)+ j sin(L

′b))

2.

(5)

To keep the width of the lines within the specificationrange of the IPD design kit, we took Zb = 35.35 � andZc = 70.7 �. With those values, the electrical length ofthe lines were reduced to Lb = 7° and Lc = 18°, whichis much smaller than the previous coupler configuration. Theamplitude imbalance was found to be equal to 0 dB at 60 GHz.However, the operating frequency was slightly shifted to lowerfrequencies. Then, by tuning again the values of La, Lb,and Lc to 34°, 10°, and 16°, respectively, we optimized theamplitude imbalance bandwidth at 1 dB for 60 GHz. However,


Fig. 9. Simulated performance of two cascaded QLQCs (crossed blue line)compared with a single QLQC (thick red line).

Fig. 10. Practical implementation of two cascaded QLQCs.

to keep a compact coupler, Lb and Lc were kept short, whichmeans that the tuning operation was mainly achieved with La

that is now 34° instead of 45°. We can observe in Fig. 12

Fig. 11. Performance of two cascaded QLQCs with a series TL (crossedblue line) compared with the performance of a single QLQC (thick red line).

that this novel coupler implementation achieves much betterperformance than all the previous configurations. Moreover,as a last tuning possibility, the capacitances C1 and C2 canhave different values if necessary.

IV. IPD INTEGRATION AT 60 GHz

A promising way to think about the integration of the ICsand the radiating element is the antenna-in-package solution.In this scheme, the antenna is realized on a low-loss substrateand can be directly integrated into the packaging. Lately,the tendency has been going to printed circuit board (PCB)package because they are low-cost. The IC is flip-chipped onthe PCB board and the antenna and beamforming network are


Fig. 12. Simulated performance of the two cascaded QLQCs with series TLand stub (crossed blue line) compared with the single QLQC (thick red line).

implemented at the RF level. Our proposed solution is to usethe IPD technology to realize the couplers and the low-losspassive circuits.

A. Novel Coupler Design and Measurement

As for the classical BLC, the signal lines of the novelcoupler were etched on the 3-μm-thick M3 layer and itsground plane on the 6-μm-thick M2 layer. The lines ofthe coupler were strongly meandered to occupy a minimum

Fig. 13. Top view of the layout of the realized novel coupler in IPD process.

space. Custom MIM capacitors between layers M2 and M3were especially optimized (Fig. 13) because the providedcapacitors from the Design Kit were only characterized atlower frequencies. Then, the ground plane was opened aroundthose MIM capacitors using a very thin 10 μm slot. The short-circuited stub was terminated by a via between M2 and M3layers.

From the designed coupler with ADS we presented before,we further miniaturized it, step by step meandering the TLs.We also had to adjust their characteristic impedance to havesuitable connections with the capacitors. The final coupleroccupies a 502 μm × 396 μm = 0.199 mm2 area (64%area reduction compared with the BLC using the same tech-nology and with the same performance, 38% with a stronglymeandered coupler with poor amplitude and PI), without theRF pads. The measured S-parameters, and amplitude and PIare shown in Fig. 14. In the 57–66 GHz band, the isolationand reflection coefficients are below −10 dB as predictedby simulation, even though their minimum are shifted downin frequency. This can be explained by the fact that thecoupler was optimized without probing pads and the measure-ments incorporate those pads which were not deembedded.The difference in amplitude and PI between simulations andmeasurements are, respectively, no more than 1.5 dB andless than 5°. We indeed achieved wideband performance. The57–65-GHz band is covered by the three −10-dB, ±1-dB, and±5° bandwidths.

B. Comparison With State-of-the-Art Published Couplers

Very few couplers were published on IPD or organic processat 60 GHz. However, a very interesting paper using capaci-tively loaded lower-ground coplanar waveguide was recentlypublished in [19]. The performance of our presented couplersis compared with this design in Table I. Our couplers achievelower insertion loss with very wideband characteristics and areduced footprint.

V. APPLICATION EXAMPLE: A BUTLER MATRIX

The use of phased-array antennas is commonly proposedto improve the budget link of a mm-wave transmission link.Compared with continuous phased arrays, discrete phasedarrays can be used if only beam steering in fixed directionsis targeted. One of the solutions consists in using the Butlermatrix. As it is a passive network, there is no power con-sumption and the complexity of the transceiver architecture


Fig. 14. Simulated (solid lines) and measured (dashed lines) S-parameters,amplitude imbalance, and PI of the novel coupler observed in Fig. 13.

Fig. 15. Schematic view of a Butler matrix.

is relaxed. As shown in Fig. 15, a 4 × 4 Butler matrix ismade of four 90° couplers, two 45° phase shifters, and twocrossovers [27].

By selecting one of the inputs (Ports 1–4) while theremaining are matched to 50 �, four different sets of phasedifferences are produced between the output ports of thematrix (Ports 5–8). Usually, the selection of the input port

Fig. 16. Layout of the fabricated Butler matrix. Ports 3, 4, 7, and 8 arematched to 50 �.

is made by an SP4T switch [28]. With each output portof the matrix feeding one antenna of the array, it thengenerates four different radiated beam angles pointing inthe ±15° and ±45° directions depending which input portis fed.

A. Butler Matrix Design and Measurement

The layout of the Butler matrix realized in IPD process isshown in Fig. 16. It uses the optimized novel 90° couplers tominiaturize its size while maintaining wideband performance.When using microstrip lines and single layer technologies,crossovers are most of the time made by cascading two 90°couplers [5], which is a quite large solution, even at 60 GHz.Here, the proposed crossovers are achieved by crossing thelines on two different layers of the IPD Process (Fig. 16,label Crossover). One of the lines remains on M3, whilethe other is laid on M2 linked by two vias. For matchingpurpose, the line on M2 is made wider. This kind of structureis very often used when multilayers technology is availablein [28]. In our case, as the ground plane is set on M2, weetched it away on purpose around the crossover. Simulatedresults, not shown here for brevity, exhibit amplitude andPIs of the crossover being, respectively, below 0.017 dB and2.8° in the whole 60-GHz band, and reflection loss higherthan 20 dB.

The 45° phase shifter is achieved by a simple delay line.Simulation results show amplitude imbalance and phase shiftbetween the microstrip lines from the crossover plus the delayline being, respectively, below 0.14 dB and 45 ±5° in thewhole 60-GHz band. The overall footprint of the Butler matrix,pads not included, is 1.6 × 1 mm2.

Since this passive network is symmetrical, only two inputports (Ports 1 and 2) and two output ports have been fed usingGSGSG probes. In the first configuration, the output GSGSGpads are connected to Ports 5 and 6 (Fig. 16), and in thesecond one, they are connected to Ports 7 and 8. The otherinput ports (Ports 3 and 4) are always matched with a 50 �load. This 50 � resistor was taken from the low frequencySTMicroelectronics DK, which is obviously not validated at60 GHz.

Fig. 17 shows the measured and simulated performance ofthe Butler matrix using the Smith chart. For a better visibility,


TABLE II

SPECIFICATIONS OF THE IMPLEMENTED BUTLER MATRIX

Fig. 17. Simulated (solid lines) and measured (dotted lines) S-parameters ofthe Butler matrix on a Smith chart when, respectively, Ports (a) 1 and (b) 2are fed (frequency ranges from 58 to 62 GHz). Phase difference between theoutput ports of the Butler matrix, respectively, when (c) Port 1 is fed andwhen (d) Port 2 is fed.

the frequency only ranges from 58 to 62 GHz. The closer weare from the center of the chart, the lower the losses of thematrix are. The dotted circle on the Smith charts represents the

Fig. 18. Performance of the proposed Butler matrix (worst insertion lossversus worst phase error) compared with the most relevant papers at 60 GHz.References [28] and [30] are 4 × 4 matrices, [29] is 8 × 8, [28] and [29] areimplemented in CMOS technology, and [30] in LTCC technology. The sizeof each circle is proportional to the area of the matrix.

minimum achievable losses with such a 4 × 4 Butler matrix,namely, 6 dB because of the power division signal in the 90°couplers. A good amplitude imbalance is obtained if all theS-parameters stand on the same circle.

This is almost true when Port 2 is fed. It can be observedin Fig. 17 that when the matrix is fed through Port 1, thesignal has a delay around 45° between the adjacent ports.When the matrix is fed through Port 2, the delay is closeto 135°. The insertion loss is lower than 11 dB at 60 GHzand also on the whole 57–66-GHz BW. If we consider theinherent loss of the hybrid couplers (6 dB since two couplersare used per path), the measured IL is then lower than 5 dB.The maximum phase error is 13.6° at 60 GHz. The reflectioncoefficient is lower than −10 dB over the 57–66-GHz bandfor all ports and the isolation is higher than 15 dB. The mainreason to explain the discrepancies between the simulationand measurements is found from the physical implementa-tion of the 50 � resistors. As we used the low frequencySTMicroelectronics Design Kit, we suspect those low fre-quency 50 � resistors to be different from pure real 50 �at 60 GHz. We estimated them to be nearly equal to 70 �.A summary of the specifications of the implemented Butlermatrix is given in Table II.

B. Comparison With State-of-the-Art Butler Matrices

Several papers present Butler matrices at 60 GHz. Fig. 18compares the performance of the proposed Butler matrix interms of insertion loss and phase error, as well as occupied size(proportional to the size of the dot) with relevant publications.


The best performance is therefore achieved with a small dotlocated in the lower left corner of the figure (see the lightarrow). Some of the structures which are presented have beenimplemented in CMOS technology [28], [29] and achieve asmaller size that the presented design. However, we achieved amatrix having much more reduced size compared with organicsubstrate design in [5], or low-temperature co-fired ceramicsin [30] with quite interesting performance. Our matrix has abetter phase error which is really important for beamformingsystems.

VI. CONCLUSION

In this paper, we have first presented a simple technolog-ical study of the IPD technology for 60-GHz applicationsthrough the design and the measurement of a TL. Then, fromthe design of an IPD BLC, we developed a novel quasi-lumped coupler with wideband characteristics to be usedover the 57–66-GHz band. This novel coupler has also areduced footprint thanks to localized capacitors, meanderedarms, and shorted stub implementation. We achieved thefollowing performance over the 57–66-GHz band: return lossand isolation higher than 10 dB, amplitude imbalance within1 dB, and PI within 5°, which is comparable or even betterthan the performance of the most recent and state-of-the-artpublished couplers at 60 GHz. This novel coupler was lastlyintegrated into a Butler matrix for RF beamforming appli-cations. Measurements showed interesting results with stillstate-of-the art performance in terms of results and occupiedspace. More important, the measurement of different couplerand Butler matrix samples showed very good reproducibilityand low variation of the IPD process making it a suitable can-didate for mm-wave passive circuits. The next step, currentlyunder study, is the association of this Butler matrix with a1 × 4 antenna array of Vivaldi antennas [31] to demonstrate afull in-package solution.

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Diane Titz (S’11–M’12) received the M.S. (Hons.)degree in telecommunications from the University ofParis-Sud (XI), and the École Normale Supérieure deCachan, Paris, France, and the Ph.D. (Hons.) degreein electrical engineering from the University of NiceSophia Antipolis, Nice, France, in 2009 and 2012,respectively.

She was with the LEAT and the CREMANT, jointlaboratories between the University of Nice andthe Orange Laboratories, France. She is currentlyan Associate Member of the EpOC team with the

University of Nice and a Full Teacher in Physics and Chemistry with the LycéeJules Ferry, Paris. She has authored and co-authored more than 10 publicationsin journals and 20 publications in international conferences. She is also areviewer for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION.Her current research interests include antenna designs, measurements, andpassive circuits, in particular, millimeter-wave frequencies.

Fabien Ferrero (M’11) was born in Nice, France,in 1980. He received the EPU Engineer degreein electronics and the master’s degree in propa-gation, télédétection, and télécommunications, andthe Ph.D. degree in electrical engineering from theUniversity of Nice-Sophia Antipolis, Nice, France,in 2003 and 2007, respectively.

He was with IMRA Europe (Aisin Seiki ResearchCenter), Valbonne, France, from 2008 to 2009, as aResearch Engineer, and developed automotive anten-nas. He is currently an Associate Professor with

the Polytechnic School, University of Nice-Sophia Antipolis. He is doinghis research with the Laboratoire d’Electronique, Antennes et Telecommuni-cations, French National Centre for Scientific Research, Paris, France, andthe Centre de Recherche Mutualisé sur les Antennes, Valbonne. His currentresearch interests include design and measurement of millimetric, miniature,and reconfigurable antennas.

Romain Pilard (S’08–M’12) received the B.S. andM.S. degrees in electronics engineering from Poly-tech’Nantes, University of Nantes, Nantes, France,and the Ph.D. degree in electrical engineering fromTelecom Bretagne, Brest, France, in 2006 and 2009,respectively.

He has been with STMicroelectronics, Crolles,France, since 2010, where he is involved in thedevelopment of integrated antennas, and high-performance passive components in advanced bulkand SOI RF CMOS technologies. His current

research interests include millimeter-wave antenna design and packagingtechnology development.

Claire Laporte received the M.S. degree inmicroelectronics from Ecole Nationale Supérieured’Electronique Informatique Radiocommunicationsde Bordeaux, Talence, France, in 2007.

She has been with STMicroelectronics, Tours,France, on the integration of passive componentson thin-film technology for RF applications, forseven years. Her current research interests includethe integration of matched baluns, directive couplers,and filtering functions for GSM, WCDMA, 4G LTE,Wifi, Bluetooth, Zigbee, and Sub-Giga applications.

Sébastien Jan received the B.S. and M.S. degreesin electronics engineering from the Institut NationalPolytechnique de Grenoble, Grenoble, France, in2004.

He has been with STMicroelectronics, Crolles,France, since 2006, where he is involved in thedevelopment of high-performance passive and activecomponents in advanced bulk and SOI RF CMOStechnologies. His current research interests includeRF, millimeter wave (noise and power), and photon-ics characterization methods.

Hilal Ezzeddine received the Ph.D. degree from theUniversity of Limoges, Limoges, France, in 2000.

He was involved with the study of the noise inthe microwave active and tunable filters. He was anAssistant Professor with the University of Limoges.Since 2001, he has been with STMicroelectronicsas an RF Designer, an RF Design Manager since2006, and in charge of the product development team(qualification and industrialization of new products)since 2012.

Frédéric Gianesello (M’13) received the B.S. andM.S. degrees in electronics engineering from theInstitut National Polytechnique de Grenoble, Greno-ble, France, and the Ph.D. degree in electrical engi-neering from Joseph Fourier University, Grenoble,in 2003 and 2006, respectively.

He is currently with STMicroelectronics, Crolles,France, where he leads the team responsible for thedevelopment of electromagnetic devices (inductor,balun, transmission line, and antenna) integratedon advanced RF CMOS/BiMOS (down to 14 nm),

silicon photonics, and advanced packaging technologies (3-D integration andfan-out wafer level packaging). He has authored and co-authored more than110 refereed journal and conference technical articles.

Dr. Gianesello has served on the TPC for the International SOI Conferencefrom 2009 to 2011, and is currently serving on the TPC for the LoughboroughAntennas and Propagation Conference.

Daniel Gloria, photograph and biography not available at the time ofpublication.


Gilles Jacquemod (M’12) received the Engineer-ing degree from the I’institut de Chimie PhysiqueIndustrielle (Chimie Physique Électronique) Lyon,Villeurbanne, France, the M.Sc. (D.E.A.) degree inmicroelectronics from Ecole Centrale Lyon, Écully,France, and the Ph.D. degree in integrated electron-ics from Institut National des Sciences Appliquéesde Lyon, Villeurbanne, France, in 1986 and 1989,respectively.

He was at LEOM, Ecole Centrale Lyon, from 1990to 2000, as an Associate Professor, on analog inte-

grated circuit design and behavioral modeling of mixed domain systems. Hewas also involved in the application of signal processing and communicationsystems. Since 2000, he has been with the LEAT Laboratory and the EcolePolytechnique of Nice-Sophia Antipolis University, Nice, France, as a FullProfessor, and with EpOC Research Team (URE UNS), since 2010. He isalso involved in RF design applied to wireless communication. He is theDirector of the CNFM PACA pole. He is the President of the CIM-PACADesign Platform. He was an Advisor to more than 25 Ph.D. students (amongwhom, four current thesis). He is the author and co-author of more than 200journal and conference papers, and holds two patents. His current researchinterests include analog integrated circuit design and behavioral modeling ofmixed domain systems.

Cyril Luxey (M’03–SM’07) was born in Nice,France, in 1971. He received the D.E.A. (Hons.)(summa cum laude) and Ph.D. (Hons.) (summa cumlaude) degrees in electrical engineering from theUniversity of Nice-Sophia Antipolis, Nice, France,in 1996 and 1999, respectively.

He was involved in several antenna-solutionsfor automotive applications like printed leaky-waveantennas, quasi-optical mixers, and retrodirectivetransponders, during his thesis. From 2000 to2002, he was with Alcatel, Mobile Phone Division,

Colombes, France, where he was involved in the design and integration ofinternal antennas for commercial mobile phones. In 2003, he was an AssociateProfessor with the Polytechnic School, University Nice Sophia-Antipolis.Since 2009, he has been a Full Professor with the IUT Réseaux et Télécoms,Sophia-Antipolis, where he is the Head of a technical bachelor degree. He isdoing his research with the EpOC Team, University of Nice-Sophia Antipolis,and is also the Co-Head of this team. In 2010, he was a Junior Member ofthe prestigious Institut Universitaire de France Institution for five years. Hiscurrent research interests include the design and measurement of millimeter-wave antennas, electrically small antennas (theoretical limits and Wheelercap techniques), antennas in-package, LTCC modules for 60- and 120-GHzapplications, multiantenna systems for diversity and MIMO techniques, andantennas for biomedical applications. He has authored and co-authored morethan 200 papers in refereed journals, international and national conferences,and book chapters. He has given more than 15 invited talks.

Dr. Luxey is an Associate Editor for the IEEE ANTENNAS AND WIRELESS

PROPAGATION LETTERS, a reviewer for the IEEE TRANSACTIONS ONANTENNAS AND PROPAGATION, the IEEE ANTENNAS AND WIRELESS

PROPAGATION LETTERS, the IEEE TRANSACTIONS ON MICROWAVE THE-ORY AND TECHNIQUES, the IEEE MICROWAVE AND WIRELESS CONFER-ENCE LETTERS, the IET Electronics Letters, the IET Microwave Antennas andPropagation journals, and several European and U.S. conferences in the fieldof microwave and antennas. He and his students received the H.W. WheelerAward of the IEEE Antennas and Propagation Society for the 2006 BestApplication Paper of the Year, the 2013 Jack Kilby Award of the IEEE Solid-State Circuits Society. He is the co-recipient of the Best Paper of the EUCAP2007 Conference, the Best Paper Award of the International Workshop onAntenna Technology (iWAT2009), the Best Student Paper at LAPC 2013 (3rdplace), and the Best Paper at LAPC 2012. He has been the General Co-Chairof the 2011 Loughborough Antennas and Propagation Conference, the awardand Grant Chair of EuCAP 2012, and the Invited Paper Co-Chair of EuCAP2013. He is a Lecturer with the European School of Antennas for the IndustrialAntennas course held at IMST. He is also one of the French Delegates of theCOST IC1102 action Versatile, Integrated, and Signal-aware Technologies forAntennas within the ICT Domain.

New Wideband Miniature Branchline Coupler on IPD Technology for Beamforming Applications

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