master in signal theory and

MASTER IN SIGNAL THEORY ANDCOMMUNICATIONS

MASTER’S THESIS

DESIGN AND CHARACTERIZATION OFW-BAND RADAR COMPONENTS

MARTA FERRERAS MAYO

2017

MASTER’S THESIS

Title: Design and Characterization of W-BandRadar Components

Author: Marta Ferreras Mayo

Tutor: Jesus Grajal de la Fuente

Department: Senales, Sistemas y Radiocomunicaciones

Group: Grupo de Microondas y Radar

COMPOSITION OF THE TRIBUNAL

President: Mariano Garcıa Otero

Vocal: Jose Manuel Riera Salıs

Secretary: Manuel Sierra Castaner

Substitute: Pedro Zufiria Zatarain

Date of defense and evaluation: 25th July 2017

Grading: 10-MH

UNIVERSIDAD POLITECNICA DE MADRID

ESCUELA TECNICA SUPERIORDE INGENIEROS DE TELECOMUNICACION

MASTER IN SIGNAL THEORY ANDCOMMUNICATIONS

MASTER’S THESIS

DESIGN AND CHARACTERIZATION OFW-BAND RADAR COMPONENTS

MARTA FERRERAS MAYO

2017

Abstract

This Master’s Thesis summarizes the work that has been performed in the frame of theSPADERADAR-CM Project for the ultimate purpose of developing a W-band space debrisradar operating at 94GHz. The main goal of this type of radars consists of detecting andtracking particles, with sizes ranging from 1 to 10 cm, that are orbiting around the Earth atspeeds up to 15 km/s and that could cause severe damage in case of collision against mannedspacecraft.

Particularly, the work performed within the realization of this Thesis has contributed to theprogress of the space debris radar in several aspects of the hardware architecture of the system.On one side, part of the work has been concerned with the characterization and integration ofthe millimeter-wave receiving subsystem of the radar. On the other side, different pre-designsfor the antenna system have been simulated and analyzed, and the performance of a reflectarray,that could be used in the future to obtain electronic scanning, has been characterized.

Apart from studying the available literature, the utilized methodology has required thefamiliarization with measurement equipment to characterize devices at millimeter wavelengths.Furthermore, the use of several high level simulation tools specialized in high frequencymodelling, such as Grasp, ADS or HFSS, has been required.

As a summary, this Master’s Thesis describes a real application of engineering, whichincludes coping with literature, designing according to specifications, simulating and performingexperimental validation through measurements.

Keywords

Radar, space debris, W-Band, millimeter-wave, quasi-optical, Cassegrain, reflectarray, Gaussianbeam, receiver, S-parameters, noise figure.

Resumen

Este Trabajo de Fin de Master expone el trabajo realizado en el marco del ProyectoSPADERADAR-CM, cuyo objetivo ultimo es el desarrollo de un radar de basura espacialembarcado que funcione a 94GHz. Este tipo de radares tienen el proposito de detectar yrealizar el seguimiento de pequenas partıculas de diametros de 1 a 10 cm que orbitan alrededorde la Tierra a velocidades del orden de 15 km/s y que pueden ocasionar graves danos en caso decolision.

En particular, este Trabajo de Fin de Master ha contribuido al avance del Proyecto endiversos aspectos de la arquitectura hardware del sistema. Por una parte, se ha caracterizado elcomportamiento lineal de la parte de milimetricas del subsistema receptor del radar. Por otraparte, se ha abordado el diseno y simulacion del sistema de antenas y se ha caracterizado elfuncionamiento de un reflectarray que, en el futuro, podrıa incorporarse al radar para conseguirexplorar el espacio mediante escaneo electronico.

Aparte del estudio de la literatura existente sobre antenas y sistemas radar, la metodologıautilizada ha requerido la familiarizacion con equipos de medida para frecuencias milimetricas.Ademas, ha sido necesario el manejo de diferentes programas de simulacion especializados en eldiseno y analisis en alta frecuencia, como son Grasp, ADS o HFSS.

Por todo ello, el trabajo expuesto en esta memoria supone un trabajo de ingenierıa real, queincorpora investigacion, diseno, simulaciones y medidas experimentales, y que por tanto, lleva ala practica muchos de los aspectos que han sido tratados en las asignaturas del Master en Teorıade la Senal y Comunicaciones.

Palabras clave

Radar, basura espacial, banda W, milimetricas, cuasi-optica, Cassegrain, reflectarray, hazgaussiano, receptor, parametros S, figura de ruido.

Contents

Abstract iii

Keywords iii

Table of contents vii

List of Figures xi

List of Tables xvii

List of Acronyms xxi

1 Introduction and Objectives 1

1.1 Motivation and Context . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.2 Objectives . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.3 Methodology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.4 Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

2 The Space Debris Radar 3

2.1 The Space Debris Problem . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.2 Spaderadar Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2.3 Spaderadar Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.3.1 Basic architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.3.2 Noise and dynamic-range considerations . . . . . . . . . . . . . . . . . . . 6

2.3.3 Monopulse radar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.3.4 Antenna system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3 Antenna System Design 9

3.1 The Spaderadar Antenna System . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

3.1.1 Cassegrain reflector system . . . . . . . . . . . . . . . . . . . . . . . . . . 10

3.1.2 Monopulse feed . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3.2 Design Criteria . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

vii

viii CONTENTS

3.2.1 Restricting dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3.2.2 Case studies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

3.3 Analytical Solution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

3.3.1 Analytical equations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

3.3.2 Analysis of the results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

3.4 Numerical Solution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

3.4.1 Simulation set-up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

3.4.2 Simulation results of different pre-designs . . . . . . . . . . . . . . . . . . 17

3.4.3 Analysis of the results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

3.5 Final Antenna Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.5.1 Geometrical definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.5.2 Simulation results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

3.5.3 Considerations on the final antenna system . . . . . . . . . . . . . . . . . 23

4 Simulation of Quasi-optical Measurement Systems 25

4.1 Theoretical Background on Quasi-Optical Systems . . . . . . . . . . . . . . . . . 25

4.1.1 Gaussian beam propagation in free space . . . . . . . . . . . . . . . . . . 25

4.1.2 Gaussian beam transformation . . . . . . . . . . . . . . . . . . . . . . . . 27

4.2 Developed Gaussian Beam Tracing Tool . . . . . . . . . . . . . . . . . . . . . . . 30

4.2.1 Running the software . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

4.2.2 Step-by-step simulation process . . . . . . . . . . . . . . . . . . . . . . . . 30

4.2.3 Simulation results and output files . . . . . . . . . . . . . . . . . . . . . . 31

4.2.4 Limitations of the simulation tool . . . . . . . . . . . . . . . . . . . . . . . 31

4.3 Application Example: 45◦ Incidence . . . . . . . . . . . . . . . . . . . . . . . . . 32

4.3.1 Design criteria . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

4.3.2 Simulated optical configurations . . . . . . . . . . . . . . . . . . . . . . . 33

4.3.3 Critical analysis of the simulation results . . . . . . . . . . . . . . . . . . 33

5 Characterization of a W-band reflectarray 35

5.1 Theoretical Background on Reflectarray Antennas . . . . . . . . . . . . . . . . . 35

5.1.1 Reflectarray antennas based on patches . . . . . . . . . . . . . . . . . . . 36

5.1.2 Reconfigurable reflectarrays based on liquid crystal . . . . . . . . . . . . . 36

5.2 Reflectarray Sample Under Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

5.3 Quasi-optical Test Benches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

5.3.1 Utilized optical components . . . . . . . . . . . . . . . . . . . . . . . . . . 39

5.3.2 Optical set-up for 30◦ incidence . . . . . . . . . . . . . . . . . . . . . . . . 39

5.3.3 Optical set-up for 45◦ incidence . . . . . . . . . . . . . . . . . . . . . . . . 40

5.3.4 Comparison of lens-based and mirror-based set-ups for 45◦ incidence . . 43

CONTENTS ix

5.4 Reflectarray Characterization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

5.4.1 Measurement plan . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

5.4.2 Static homogeneous control . . . . . . . . . . . . . . . . . . . . . . . . . . 46

5.4.3 Dynamic control based on time-multiplexing . . . . . . . . . . . . . . . . 51

5.4.4 Discussion of the results . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

5.4.5 Future measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

6 Radar Receiving Chain Characterization. 57

6.1 The Millimeter-Wave Receiving Subsystem . . . . . . . . . . . . . . . . . . . . . 57

6.2 Characterization of Individual Components . . . . . . . . . . . . . . . . . . . . . 58

6.3 Characterization of the Receiver Isolation Chain . . . . . . . . . . . . . . . . . . 59

6.3.1 Transmit insertion losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . 60

6.3.2 Isolation between the transmitter and the receiver. . . . . . . . . . . . . . 60

6.3.3 Power transfer from the antenna to the receiver. . . . . . . . . . . . . . . 62

6.4 Noise Performance of the Receiver Chain . . . . . . . . . . . . . . . . . . . . . . 64

6.4.1 Analytical estimation using Friis formula . . . . . . . . . . . . . . . . . . 65

6.4.2 Noise budget analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

6.4.3 Noise measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

6.4.4 Conversion losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

6.5 Overall Conclusions from the Measurements . . . . . . . . . . . . . . . . . . . . . 71

6.5.1 Transmit-receive isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

6.5.2 Maximum output power of the transmitter . . . . . . . . . . . . . . . . . 72

6.5.3 Receiver noise floor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

6.5.4 Receiver sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

7 Summary and Conclusions 75

7.1 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

7.2 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

A Simulations in Grasp 77

A.1 PO and PTD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77

A.2 Grasp configuration for simulating a Cassegrain system . . . . . . . . . . . . . . 77

A.2.1 Cassegrain antenna model . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

A.2.2 Command list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

B Optical Test Benches for 45◦ Incidence 81

B.1 Optical Set-up Using Two Dielectric Lenses . . . . . . . . . . . . . . . . . . . . . 81

B.2 Optical Set-up Using Two 45◦ Off-axis Mirrors . . . . . . . . . . . . . . . . . . . 83

B.2.1 Option with available 45◦ off-axis mirrors . . . . . . . . . . . . . . . . . . 83

x CONTENTS

B.2.2 Option with alternative 45◦ off-axis mirrors . . . . . . . . . . . . . . . . . 84

B.3 Optical Set-up Using Two 90◦ Off-axis Mirrors . . . . . . . . . . . . . . . . . . . 85

B.4 Optical Set-up Using 45◦ Off-axis Mirrors and Lenses . . . . . . . . . . . . . . . 87

B.5 Optical Set-up Using 45◦ and 90◦ Off-axis Mirrors . . . . . . . . . . . . . . . . . 89

B.5.1 Results for option 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

B.5.2 Results for option 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92

C Phase Center Calculation 93

C.1 RPG FH-PP-100 Potter Horn . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93

C.2 Millitech SGH-08 Conical Horn . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96

D Measurement Set-ups and Equipment Configurations 99

D.1 S-Parameters Characterization of W-band Devices . . . . . . . . . . . . . . . . . 99

D.2 Noise Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101

D.3 Free-Space Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102

E Free-space Post-processing Techniques 105

E.1 Recalibration with a Reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105

E.2 Time-domain Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

E.3 Smoothing vs Time-domain Processing . . . . . . . . . . . . . . . . . . . . . . . . 107

F Characterization of Millimeter-wave Components 109

F.1 RPG WFI-110 Isolator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 110

F.2 RPG WPD-110 Hybrid Power Divider . . . . . . . . . . . . . . . . . . . . . . . . 111

F.3 ELVA CR-1094 Circulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114

F.4 Quinstar QAL-W00000 Variable Attenuator . . . . . . . . . . . . . . . . . . . . . 116

F.5 ELVA SPST-10 Switch . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118

F.6 Joint Response of the Isolator and the Switch . . . . . . . . . . . . . . . . . . . . 120

F.7 RPG W-LNA75110 Low Noise Amplifier . . . . . . . . . . . . . . . . . . . . . . . 123

G Millimeter Wave Components Datasheets 125

Bibliography 141

List of Figures

2.1 Block diagram of the preliminary radar architecture. . . . . . . . . . . . . . . . . 5

2.2 Modified block diagram of the radar architecture using and two channels formonopulse tracking. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

2.3 Classical Cassegrain antenna schematic. . . . . . . . . . . . . . . . . . . . . . . . 7

3.1 Centred Cassegrain antenna geometry. . . . . . . . . . . . . . . . . . . . . . . . . 10

3.2 Gain curves with respect to the main reflector diameter. . . . . . . . . . . . . . . 12

3.3 Position and approximate dimensions of the monopulse horn. . . . . . . . . . . . 12

3.4 Feed directivity model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

3.5 Analyzed Cassegrain geometries from Table 3.2. . . . . . . . . . . . . . . . . . . . 17

3.6 Simulated farfield results for geometrical design 1. . . . . . . . . . . . . . . . . . 17




3.10 Geometry of the final design of the dual antenna system. . . . . . . . . . . . . . . 22

3.11 Simulated farfield of the final design illuminated by C = −10 dB. . . . . . . . . . 23

4.1 Evolution of the beam radius and phase radius of curvature for a Gaussian beamthat propagates in a certain direction. . . . . . . . . . . . . . . . . . . . . . . . . 26

4.2 Gaussian beam transformation by lens. . . . . . . . . . . . . . . . . . . . . . . . . 28

4.3 Example of an afocal system for quasi-optical beam propagation. . . . . . . . . . 29

5.1 Schematic of a reflectarray antenna. . . . . . . . . . . . . . . . . . . . . . . . . . 36

5.2 Schematic of the manufactured reflectarray. . . . . . . . . . . . . . . . . . . . . . 37

5.3 Photographs of the reflectarray sample. . . . . . . . . . . . . . . . . . . . . . . . 38

5.4 Test bench configuration for 30◦ incidence at 94GHz. . . . . . . . . . . . . . . . 40

5.5 Test bench configuration for 45◦ incidence at 94GHz using lenses. . . . . . . . . 41

5.6 Test bench configuration for 45◦ incidence at 94GHz using mirrors. . . . . . . . . 42

5.7 Optical thru in the quasi-optical test bench utilized to measure vertical incidenceresponse for 45◦ incidence. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43

5.8 Measured S21 of the measured optical thru for 45◦ incidence. . . . . . . . . . . . 44

xi

xii LIST OF FIGURES

5.9 Reflectarray in the quasi-optical test bench utilized to measure vertical orientationresponse for 30◦ incidence. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

5.10 Reflectarray measurements using sinusoidal excitation, 30◦ incidence and verticalorientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

5.11 Reflectarray measurements using square excitation for 30◦ incidence and verticalorientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47


5.13 Reflectarray measurements using sinusoidal excitation, 30◦ incidence andhorizontal orientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

5.14 Reflectarray measurements using square excitation for 30◦ incidence andhorizontal orientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48


5.16 Reflectarray measurements using sinusoidal excitation for 45◦ incidence andvertical orientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

5.17 Reflectarray measurements using square excitation for 45◦ incidence and verticalorientation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

5.18 Comparison of the measured phase ranges at 94GHz and 100GHz. . . . . . . . . 50

5.19 Comparison of the measured losses at 94GHz and 100GHz. . . . . . . . . . . . . 50

5.20 Reflectarray performance for the four analyzed dynamic states for verticalorientation: strategy 1. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

5.21 Reflectarray performance for the four analyzed dynamic states for horizontalorientation: strategy 1. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

5.22 Reflectarray performance for the four analyzed dynamic states for horizontalorientation: strategy 2. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

5.23 Mean values of the phase curves for the different dynamic excitation strategies. . 55

6.1 Millimeter-wave receiving subsystem options. . . . . . . . . . . . . . . . . . . . . 58

6.2 Important situations of the receiver that need to be analyzed. . . . . . . . . . . . 60

6.3 Measurement set-up for the isolation chain collocating the load at port 2 of thecirculator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

6.4 Isolation chain performance from the transmitter to the receiver: S-parameters. . 61

6.5 Isolation chain performance from the transmitter to the receiver: insertion andreturn losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

6.6 Measurement set-up for the isolation chain collocating the load at port 1 of thecirculator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

6.7 Isolation chain performance from the antenna port to the LNA port: S-parameters. 63

6.8 Isolation chain performance from the antenna port to the LNA port: insertionand return losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

6.9 Block diagram of the receiver. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

6.10 Receiver schematic for the ADS noise budget analysis. . . . . . . . . . . . . . . . 67

LIST OF FIGURES xiii

6.11 Noise budget analysis of the receiver. . . . . . . . . . . . . . . . . . . . . . . . . . 68

6.12 Measurement set-up for the characterization of the receiver chain noise performance. 69

6.13 Noise power measurement at the output of the receiver chain when the transmitterand the antenna are matched (Ta = 290K). . . . . . . . . . . . . . . . . . . . . . 69

6.14 Impact of using different input power values to the multiplier. . . . . . . . . . . . 70

6.15 Schematic of the isolation chain. . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

B.1 Proposed optical arrangement using horns and lenses. . . . . . . . . . . . . . . . 82

B.2 Simulation results for the sample size that obtains �17.4 dB taper using twodielectric lenses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82

B.3 Proposed optical set-up using horns and 45◦ off-axis mirrors. . . . . . . . . . . . 83

B.4 Simulation results for the sample size that obtains �17.4 dB taper using 45◦

off-axis mirrors with 101.6mm diameter . . . . . . . . . . . . . . . . . . . . . . . 84


off-axis mirrors with 76.2mm diameter. . . . . . . . . . . . . . . . . . . . . . . . 85

B.6 Proposed optical arrangements using of horns and 90◦ off-axis mirrors to measurereflectivity at 45◦ incidence. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86


off-axis mirrors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86

B.8 Proposed optical set-up using of horns, 45◦ off-axis mirrors and lenses. . . . . . . 88

B.9 Simulation results for the sample size that obtains �17.4 dB taper using two lensesand two 45◦ off-axis mirrors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

B.10 Proposed optical arrangements using of horns, 45◦ and 90◦ off-axis mirrors. . . . 90

B.11 Simulation results for the sample size that obtains �17.4 dB taper using 45◦ and90◦ off-axis mirrors (option 1). . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91

B.12 Simulation results for the sample size that obtains �17.4 dB taper using 45◦ and90◦ off-axis mirrors (option 2). . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92

C.1 HFSS model of the RPG horn using symmetries. . . . . . . . . . . . . . . . . . . 94

C.2 Phase of the copolar component of the electric field at 94GHz when the coordinatesystem is at z = 20.5mm inside the horn. . . . . . . . . . . . . . . . . . . . . . . 94

C.3 Phase of the copolar component of the electric field at 100GHz when thecoordinate system is at z = 24.5mm inside the horn. . . . . . . . . . . . . . . . . 95

C.4 Directivity at 94GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95


C.6 HFSS model of the Millitech horn using symmetries. . . . . . . . . . . . . . . . . 96

C.7 Phase of the copolar component of the electric field at 94GHz when the coordinatesystem is at z = 6.6mm inside the horn. . . . . . . . . . . . . . . . . . . . . . . . 97

C.8 Phase of the copolar component of the electric field at 100GHz when thecoordinate system is at z = 7.9mm inside the horn. . . . . . . . . . . . . . . . . . 97


xiv LIST OF FIGURES


D.1 Measurement set-up used for the acquisition of the S-parameters ofmillimeter-wave devices. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100

D.2 Measurement of a Quinstar W-band load. . . . . . . . . . . . . . . . . . . . . . . 101

D.3 Measurement set-up for the acquisition of the noise power level at the output ofthe receiving chain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102

D.4 Coaxial cable and WR-10 transitions between the horns and the VNA (VectorNetwork Analyzer) ports. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103

D.5 Thru of the TRL calibration kit after calibration. . . . . . . . . . . . . . . . . . . 104

E.1 Photograph of the set-up to measure the optical thru. . . . . . . . . . . . . . . . 105

E.2 Calibration of the transmission coefficient with an optical thru. . . . . . . . . . . 106

E.3 Time domain transformation of the S-parameters of the 30◦-incidence optical testbench. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107

E.4 Post-processing of the transmission coefficient. . . . . . . . . . . . . . . . . . . . 108

F.1 Measurement set-up for the RPG WFI-110 isolator. . . . . . . . . . . . . . . . . 110

F.2 RPG WFI-110 isolator: S-parameters. . . . . . . . . . . . . . . . . . . . . . . . . 110

F.3 RPG WFI-110 isolator: other measurements. . . . . . . . . . . . . . . . . . . . . 111

F.4 Measurement set-up for the RPG WPD-110 power divider. . . . . . . . . . . . . 112

F.5 RPG WPD-110 hybrid power divider: S-parameters. . . . . . . . . . . . . . . . . 112

F.6 RPG WPD-110 hybrid power divider: insertion and return losses. . . . . . . . . . 113

F.7 RPG WPD-110 hybrid power divider: phase balanced and group delay. . . . . . 113

F.8 Measurement set-up for the ELVA CR-1094 circulator. . . . . . . . . . . . . . . . 114

F.9 ELVA CR-1094 circulator: S-parameters. . . . . . . . . . . . . . . . . . . . . . . . 115

F.10 ELVA CR-1094 circulator: other measurements. . . . . . . . . . . . . . . . . . . . 116

F.11 Measurement set-up for the Quinstar QAL-W00000 variable attenuator. . . . . . 116

F.12 Quinstar QAL-W00000 attenuator: return losses. . . . . . . . . . . . . . . . . . . 117

F.13 Quinstar QAL-W00000 attenuator: attenuation and phase shift. . . . . . . . . . 117

F.14 Measurement set-up for the ELVA SPST-10 switch. . . . . . . . . . . . . . . . . . 118

F.15 ELVA SPST-10 switch: S-parameters. . . . . . . . . . . . . . . . . . . . . . . . . 119

F.16 ELVA SPST-10 switch: insertion losses. . . . . . . . . . . . . . . . . . . . . . . . 119

F.17 ELVA SPST-10 switch: return losses. . . . . . . . . . . . . . . . . . . . . . . . . . 120

F.18 Measurement set-up to characterize the joint response of the isolator and the switch.121

F.19 Isolator + switch: S-parameters. . . . . . . . . . . . . . . . . . . . . . . . . . . . 121

F.20 Isolator + switch: return losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122

F.21 Isolator + switch: insertion losses. . . . . . . . . . . . . . . . . . . . . . . . . . . 122

F.22 Measurement set-up for the RPG W-LNA75110 low noise amplifier. . . . . . . . 123

F.23 RPG W-LNA75110 low noise amplifier: S-parameters. . . . . . . . . . . . . . . . 123

LIST OF FIGURES xv

F.24 RPG W-LNA75110 low noise amplifier: gain and return losses. . . . . . . . . . . 124

xvi LIST OF FIGURES

List of Tables

2.1 Preliminary operational parameters of the radar system. . . . . . . . . . . . . . . 4

2.2 Radar capabilities expected from the pre-design values. . . . . . . . . . . . . . . 4

3.1 General specifications of the radar antenna system. . . . . . . . . . . . . . . . . . 9

3.2 Study cases for the Cassegrain antenna system. . . . . . . . . . . . . . . . . . . . 14

3.3 Analytical results for different specifications. . . . . . . . . . . . . . . . . . . . . . 15

3.4 Gaussian feed configuration for each of the four compared geometries so thatillumination at the edge of the subreflector is �10 dB. . . . . . . . . . . . . . . . 16

3.5 Comparison of the results obtained for different pre-designs. . . . . . . . . . . . . 19

3.6 Geometrical and simulated results of the chosen configuration. . . . . . . . . . . 21

3.7 Geometrical parameters of the final Cassegrain system. . . . . . . . . . . . . . . . 22

3.8 Simulation results for the final Cassegrain design optimally illuminated by aGaussian feed. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

4.1 Recommended sample sizes for using different free-space measurement systems(using RPG horns). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

5.1 Specifications of the reflectarray sample. . . . . . . . . . . . . . . . . . . . . . . . 37

5.2 Element positions for the proposed optical test bench for 30◦ incidence. . . . . . 40

5.3 Simulated results of the proposed test bench configuration to measure thereflectarray with an angle of incidence of 30◦. . . . . . . . . . . . . . . . . . . . . 40

5.4 Element positions for the proposed optical test bench for 45◦ incidence using lenses. 41

5.5 Simulated results of the test bench configuration to measure the reflectarray withan angle of incidence of 45◦ using lenses. . . . . . . . . . . . . . . . . . . . . . . . 42

5.6 Element positions for the proposed optical test bench for 45◦ incidence usingmirrors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

5.7 Simulated results of the alternative test bench configuration to measure thereflectarray with an angle of incidence of 45◦ using mirrors. . . . . . . . . . . . . 43

5.8 Common parameters to all the measurements. . . . . . . . . . . . . . . . . . . . . 45

5.9 Summary of both excitation strategies for dynamic time-multiplexed control ofthe reflectarray states. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

5.10 Summary of the reflectarray performance when time-multiplexing is used. . . . . 56

xvii

xviii LIST OF TABLES

6.1 Summary of the measured responses of the receiving chain millimeter-wavecomponents at 94GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

6.2 Transmitter-to-antenna performance at 94GHz. . . . . . . . . . . . . . . . . . . . 60

6.3 Transmitter-to-receiver performance results at 94GHz. . . . . . . . . . . . . . . . 62

6.4 Antenna-to-receiver performance results at 94GHz. . . . . . . . . . . . . . . . . . 64

6.5 Friis formula parameters at 94GHz. . . . . . . . . . . . . . . . . . . . . . . . . . 66

6.6 Output noise results at 94GHz for different receiver bandwidths obtained usingFriis formula. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

6.7 Noise budget analysis results at 94GHz at the output of the IF (IntermediateFrequency) active filter for different receiver bandwidths. . . . . . . . . . . . . . . 68

6.8 Noise power level at the output of the receiver chain for an antenna temperatureof Ta ≈ Tamb ≈ 290K. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

6.9 Most relevant measurement results of the receiving chain at 94GHz. . . . . . . . 71

6.10 Comparison of noise power results at the output of the receiver chain accordingto analytical calculations, simulations and measurements. . . . . . . . . . . . . . 72

6.11 Receiver sensitivity for a swerling 5 target without pulse integration and Ta = 290K. 73

A.1 Textual reproduction of the Grasp command list. . . . . . . . . . . . . . . . . . . 78

B.1 Element positions for the proposed arrangement using horns and lensese. . . . . 82

B.2 Results at 94GHz and 100GHz obtained using a configuration that utilizes hornsand dielectric lenses for 45◦ incidence. . . . . . . . . . . . . . . . . . . . . . . . . 83

B.3 Element positions for the proposed set-up using horns and 45◦ off-axis mirrors. . 83

B.4 Results at 94GHz and 100GHz obtained using a configuration that uses hornsand 45◦ off-axis mirrors with 101.6mm diameter. . . . . . . . . . . . . . . . . . . 84

B.5 Results at 94GHz and 100GHz obtained using a configuration that uses hornsand 45◦ off-axis mirrors with 76.2mm diameter. . . . . . . . . . . . . . . . . . . . 85

B.6 Element positions for the proposed arrangement (option 2) using horns and 90◦

off-axis mirrors to measure reflectivity at 45◦ incidence. . . . . . . . . . . . . . . 86

B.7 Results at 94GHz and 100GHz obtained using a configuration that uses hornsand 90◦ off-axis mirrors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87

B.8 Element positions for the proposed set-up using horns, 45◦ off-axis mirrors andlenses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

B.9 Minimum sample diameter at 94GHz and 100GHz obtained using a configurationthat utilizes horns and dielectric lenses. . . . . . . . . . . . . . . . . . . . . . . . 89

B.10 Element positions for the proposed arrangement using horns, 45◦ and 90◦ off-axismirrors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

B.11 Results at 94GHz and 100GHz obtained using a configuration that uses horns,45◦ and 90◦ off-axis mirrors (option 1). . . . . . . . . . . . . . . . . . . . . . . . . 90

B.12 Results at 94GHz and 100GHz obtained using a configuration that uses horns,45◦ and 90◦ off-axis mirrors (option 2). . . . . . . . . . . . . . . . . . . . . . . . . 92

LIST OF TABLES xix

C.1 Phase center at different frequencies measured from the aperture (z > 0 is insidethe horn). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94

C.2 Phase center at different frequencies measured from the aperture (z > 0 is insidethe horn). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96

D.1 Anritsu VNA configuration for the measurement of S-parameters at W-band. . . 100

D.2 Agilent spectrum analyzer configuration for the measurement of noise level of thereceiving chain. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102

D.3 Anritsu VNA configuration for the free-space measurements at W-band. . . . . . 103

F.1 RPG WFI-110 isolator measurement results at 94GHz in comparison with thoseprovided by the manufacturer. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111

F.2 RPG WPD-110 hybrid power divider measurement results at 94GHz incomparison with those provided by the manufacturer. . . . . . . . . . . . . . . . 114

F.3 ELVA CR-1094 circulator measurement results at 94GHz in comparison withthose provided by the manufacturer. . . . . . . . . . . . . . . . . . . . . . . . . . 115

F.4 Quinstar QAL-W00000 attenuator measurement results at 94GHz in comparisonwith those provided by the manufacturer. . . . . . . . . . . . . . . . . . . . . . . 118

F.5 ELVA SPST-10 switch: biasing and power consumption. . . . . . . . . . . . . . . 118

F.6 ELVA SPST-10 switch measurement results at 94GHz in comparison with thoseprovided by the manufacturer. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120

F.7 Measurement results of the switch and the isolator at 94GHz. . . . . . . . . . . . 123

F.8 RPG W-LNA75110 low noise amplifier measurement results at 94GHz incomparison with those provided by the manufacturer. . . . . . . . . . . . . . . . 124

xx LIST OF TABLES

List of Acronyms

ADC Analog Digital Converter

ADS Advanced Design System R©

DDS Direct Digital Synthesis

DSB Double-Side Band

DUT Device Under Test

FEM Finite Element Method

FFT Fast Fourier Transform

GEA Applied Electromagnetism Group

GMR Microwave and Radar Group

GR Radiation Group

Grasp General Reflector and Antenna Farm Analysis Software R©

HFSS High Frequency Structural Simulator R©

HPA High Power Amplifier

IF Intermediate Frequency

LC Liquid Crystal

LCD Liquid Crystal Display

LFM Linear Frequency Modulation

LHCP Left-Hand Circular Polarization

LNA Low Noise Amplifier

LO local oscillator

MMIC Monolithic Microwave Integrated Circuit

OMT Orthomode Transducer

xxi

xxii List of Acronyms

PLO Phased Locked Oscillator

PO Physical Optics

PRI Pulse Repetition Interval

PTD Physical Theory of Diffraction

PW Pulse Width

RBW Resolution Bandwidth

RF Radiofrequency

RHCP Right-Hand Circular Polarization

RMS Root Mean Square

SNR Signal to Noise Ratio

SSB Single-Side Band

TRL Through-Reflect-Line

UPM Technical University of Madrid

VBW Video Bandwidth

VNA Vector Network Analyzer

VSWR Voltage Standing Wave Ratio

Chapter 1

Introduction and Objectives

1.1 Motivation and Context

In the frame of the SPADERADAR-CM Project [1, 2, 3], a W-band space debris radar is beingdeveloped by a consortium lead by the GMR (Microwave and Radar Group) of UPM (TechnicalUniversity of Madrid). This type of radars must be capable of detecting targets with sizes inthe range of 1 to 10 cm that are orbiting around the Earth at speeds of 15 km/s and that cancause severe damage in case of collision [4].

Particularly, the Spaderadar Project is focused on the development of spaceborne radars,which are able to detect and track tiny particles that approximate dangerously to theinfrastructure that is being protected. Therefore, their angular resolution, maximum rangeand frame rate must be sufficiently high to produce real-time responses to incoming threats [2].

This Master’s Thesis summarizes the work that has been performed in the GMR for thepurpose of developing, characterizing and integrating part of the millimeter-wave receivingsubsystem of the space debris radar that is being deployed.

1.2 Objectives

This Master’s Thesis will be concerned with three different tasks related to the space debrisradar:

1. The first task will consist of making a proposal for the antenna system that will beintegrated in the radar prototype. The basic system will take the form of a Cassegrainantenna fed by a monopulse horn. In particular, the geometrical definition of theCassegrain system will be tackled taking into account the application requirements.

2. In order to avoid the mechanical scanning that is inherent to the basic explorationapproach, it has been considered the substitution of the hyperbolic subreflector bya reflectarray antenna. To this end, the second task of this Thesis will consist ofcharacterizing a reflectarray (already manufactured) at W-band. As a preliminary step, itwill be necessary to design an optical bench set-up for free-space reflectivity measurements,for which a software based on Gaussian beam propagation has been developed.

3. The last goal of this Thesis is the characterization of the components of one of the receivingchannels of the radar. The intended characterization includes the acquisition of some of

1

2 1.3. METHODOLOGY

the most representative figures of merit of electronic devices, such as the S-parameters orthe noise factor.

1.3 Methodology

The methodology that has been followed is similar for all the tasks described in Section 1.2:

1. Documentation on the subject by investigating the available literature.

2. Design and/or simulation using high level software tools specialized for high frequencydesigns, such as Grasp, ADS or HFSS. Matlab has been used for the development of theGaussian beam tracing tool.

3. Realization of measurements using adequate equipment for characterizing high frequencydevices. Processing and presentation of the results are performed using Matlab.

4. Critical analysis of the obtained results, both from simulations and from measurements,in comparison to what it was specified or simulated.

1.4 Structure

From this point, the document is divided into six chapters:

• Chapter 2 offers an overview of the general specifications and topology of the SpaderadarProject emphasizing those aspects that will be developed along the rest of the Thesis.

• Chapter 3 describes the design procedure of the dual reflector system, the results fromdifferent iterations and the different trade-offs that had to be balanced in order to obtainthe final pre-design of the Cassegrain antenna system.

• Chapter 4 describes the work realized in relation to the development of a software toolbased on Gaussian Beam Theory to predict the behaviour of waves in quasi-optical systems.

• Chapter 5 summarizes the results that have been obtained after characterizing areflectarray at W-band using a free-space measurement system.

• Chapter 6 summarizes the results that have been obtained after the characterization ofone of the channels of the millimeter-wave receiving subsystem of the radar.

• Chapter 7 presents some general conclusions about the work performed within therealization of this Thesis.

Appendices A through G compliment the information included along these chapters.

Chapter 2

The Space Debris Radar

This chapter describes the specifications, functionality and architecture of the space debris radarthat is being developed in the frame of the SPADERADAR-CM Project [1, 2] and it introducessome of the tasks that will be covered along the rest of the document.

2.1 The Space Debris Problem

Space debris consists of micrometeoroids (natural) and remnants of spacecraft and vehicles(man-made). The growing number of spaceborn missions and applications has produced tenthsof millions of particles that are polluting the orbital environment[4].

Since the hazards associated with space debris are continuously increasing, it is only amatter of time until a manned system is hit with potentially catastrophic consequences. Indeed,collisions with space debris are a reality of space flight today. Therefore, risks associated to suchcollisions must be managed, which is becoming a crucial task for the national and internationalspace agencies [4].

Two different strategies exist to combat orbital debris:

• Protection through debris shielding.

• Avoidance through debris detection.

New spacecraft now incorporate debris shields. However, debris protection is limited tosmall particles (< 1 cm) and old spacecraft are still vulnerable. Therefore, attention is alsobeing paid to the deployment of early detection systems that can produce a warning long beforethe collision.

Radar systems have a fundamental role in the observation of space debris. However, currentlydeployed terrestrial radars are not able to detect and track objects with sizes below 10 cm [5].As an alternative or complement to the surveillance capabilities provided by terrestrial radars,spaceborne radars may be used:

• They do not suffer from atmospheric attenuation, so millimeter waves can be utilized toincrease resolution up to 0.1 to 1 cm.

• They are closer to the targets, so they require less transmitted power and smaller antennas.

3

4 2.2. SPADERADAR SPECIFICATIONS

• Real-time processing of the acquired data could trigger autonomous threat-avoidanceprocesses in case particles were approaching dangerously to the platform that is beingprotected .

[5, 6] summarize the characteristics of some of the currently deployed radars which would becapable of space debris detection.

2.2 Spaderadar Specifications

The ultimate purpose of the SPADERADAR-CM Project is to develop a W-band space debrisradar operating at 94GHz [1, 2, 3]. The projected system will be a prototype of a spaceborneradar that must be capable of detecting targets with sizes in the range of 1 to 10 cm that areorbiting around the Earth at speeds up to 15 km/s and that can cause severe damage in case ofcollision with any manned spacecraft [2].

The preliminary design of the space debris radar is described in [2]. That document concludesthat the radar system will be a pulse radar using LFM (Linear Frequency Modulation) pulses andworking at 94GHz. The preliminary operative parameters of the radar system are summarizedin Table 2.1. Those values might vary in case any limitations are found along the implementationprocess.

Parameter Value

Operating frequency 94GHz

Transmitted bandwidth ≈ 50MHz

Antenna gain > 55 dB

Transmitted power 30 dBm [1]

PRI (Pulse Repetition Interval) 100 �s [1]

PW (Pulse Width) 24 �s [1]

Sampling frequency 100MHz

Table 2.1: Preliminary operational parameters of the radar system.

Those parameters would yield the system-level capabilities presented in Table 2.2 [2].

Parameter Value

Maximum range (d=10 cm, SNR=13 dB) 10.82 km

Maximum Doppler frequency 10MHz

Spatial resolution 3m

Angular resolution 0.3◦

Table 2.2: Radar capabilities expected from the pre-design values.

[1]These parameters are yet to be decided.

CHAPTER 2. THE SPACE DEBRIS RADAR 5

2.3 Spaderadar Architecture

2.3.1 Basic architecture

Figure 2.1 shows a block diagram of the proposed pulsed LFM radar architecture using anheterodyne receiver.

PLO3.875 GHz

DDS

CLK

PC

Master Oscillator

(10 MHz)

Microwave

SubsystemDigital

Electronics

CH1

Generador

señal

CTL

Filter 1

PLO3 GHz

x4 MPA

Filter 2

x4

Filter 3

LO (15,5 GHz)

LO2 (3.875 GHz)

3 GHz

MPA

Pulsed LFM Signal

(916.667- 918.7 MHz )

Pulsed LFM Signal

(15.6667 – 15.675 GHz ) Pulsed LFM Signal

(3.916667- 3.9187 GHz )

LNA

MPA HPA

x6

Millimiter – wave

Subsystem

x6

IF (0-1 GHz)

Pulsed LFM Signal

(94 -94.05 GHz )

LO (93-94 GHz)

IQ Receiver

Subsystem

I Samples

Q Samples

Transmitter/

Receiver

Antenna

Figure 2.1: Block diagram of the preliminary radar architecture [3].

Digital electronics: DDS (Direct Digital Synthesis) is a technique to generate LFM signals.In this case, sawtooth pulsed chirp signals of 50MHz are generated every 100 �s at 917MHzusing a commercial DDS.

Microwave subsystem: The microwave subsystem comprises the RF (Radiofrequency)circuitry up to 20GHz. Components are based on planar technologies and they canincorporate commercial MMIC (Monolithic Microwave Integrated Circuit) resulting inlow-cost compact designs.

A 3GHz PLO (Phased Locked Oscillator) and a ×4 multiplier upconvert the signal andthe LO (local oscillator) to 15.67GHz.

Millimeter-wave subsystem: The millimeter-wave subsystem involves the componentsoperating above 20GHz, which are usually based on waveguide technology. The goalsof this subsystem are frequency multiplication (×6) and amplification of the pulsed signal.

The receiver utilizes a mixer to downconvert the received pulses to an IF of 0 - 1GHz (tobe decided).

IQ receiver subsystem: Its principal task is to transform the IF signal into IQ samples, eitherin analog or digital domain.

Antenna system: The antenna system will consist of a dual reflector antenna with a gain of55 - 60 dB. The simplest system utilizes a mechanical scanning approach to explore thespace.

In the previous design, the LO frequency is a pure tone that downconverts the RF signal toan IF that can be digitalized by the ADC (Analog Digital Converter). Signal pulse compressionis then performed entirely in the digital domain.

6 2.3. SPADERADAR ARCHITECTURE

2.3.2 Noise and dynamic-range considerations

Receivers generate internal noise that can mask weak signals. The strategy to maintain a lownoise figure consists of introducing an LNA (Low Noise Amplifier) at the input of the receivingchain, so that this LNA establishes the noise floor of the receiver [7].

However, since this is a monostatic radar, two problems may arise: the high transmittedpower reaching the receiver through the circulator or after reflecting at the antenna, and thereceived power being too high due to early targets. To avoid damages on the receiver components,the receiver must remain switched-off during the time the transmitter is active. For this purpose,an isolation chain formed by a circulator, an isolator and a switch is introduced before the LNA.This will degrade the overall noise figure, since the LNA no longer is the first component of thereceiver [8].

On the other hand, the noise introduced by the ADC is usually analyzed as a separatecontribution to the overall radar noise. If the quantification noise introduced by the ADC ishigher than the receiver noise power, low energy echoes could be obscured, thus reducing thedynamic range of the system [7]. In order to ensure that the receiver dynamic range is limitedby thermal noise, an IF active filter has been included after the mixer.

All these problems will be analyzed in Chapter 6 after characterizing the receiverperformance.

2.3.3 Monopulse radar

Monopulse tracking is based on the minimization of an error signal that is dependent on thetarget displacement from the pointing axis [7]. For this radar, a multi-mode horn is beingdesigned along with a waveguide mode extractor, allowing a much flexible design than amulti-horn structure. This design is not part of the work included in this Thesis.

A monopulse system based on amplitude comparison would require three different receivingchannels to provide tracking error signals for two dimensions: sum, elevation-difference andazimuth-difference. On the other hand, if the system is designed to use circular polarization,it will be possible to obtain azimuth and elevation control using the sum pattern and a singledifference signal [9].

The introduction of monopulse tracking requires modifications on the receive subsystems ofthe radar architecture. Basically, the new architecture must duplicate or triplicate (dependingon the number of channels that are finally utilized) the receiving chains, so that every outputsignal of the monopulse comparator is downconverted and digitalized. The new architecture isshown in Figure 2.2.

CHAPTER 2. THE SPACE DEBRIS RADAR 7

PLO3.875 GHz

DDS

CLK

PC

Master Oscillator

(10 MHz)

Microwave

SubsystemDigital

Electronics

CH1

Generador

señal

CTL

Filter 1

PLO3 GHz

x4 MPA

Filter 2

x4

Filter 3

LO (15,5 GHz)

LO (3.875 GHz)

3 GHz

MPA

Pulsed LFM Signal

(916.667- 918.7 MHz )

Pulsed LFM Signal

(15.6667 – 15.675 GHz ) Pulsed LFM Signal

(3.916667- 3.9187 GHz )

LNA

MPA HPA

x6

Millimiter – wave

Subsystem

x6

IF (0-1 GHz)

Pulsed LFM Signal

(94 -94.05 GHz )

LO (93 GHz)

Transmitter/

Receiver

Antenna

LNA

Mono

pulse

HMC370LP4 HMC451LP1

Synergy

LFSN200400-100

Synergy

LFSN200400-100

Farran

FPA-10-0002Farran

FPA-10-0005

RPG

AFM6-110

RPG

AFM6-110

Quinstar

QMB-9999WS

RPG

WLNA-75-110

Elva

Switch

HMC451LP1

RPG

WLNA-75-110Quinstar

QMB-9999WS

Elva

Switch

IF (1-1.04 GHz)IQ Receiver

Subsystem

I Samples

Q Samples

IQ Receiver

Subsystem

I Samples

Q Samples

Figure 2.2: Modified block diagram of the radar architecture using andtwo channels for monopulse tracking [3].

2.3.4 Antenna system

The application requires an antenna with more than 55 dB gain at W-band that can performthree-dimensional scanning. Besides, this is a spaceborne application, so it is also considerablyimportant that the entire antenna system is as compact as possible. Considering this, the spacedebris radar will utilize an axially symmetric Cassegrain configuration [10] whose schematic ispresented in Figure 2.3. Some pre-designs of this reflector antenna system are analyzed alongChapter 3.

F2F1

Main

reflector

SubreflectorFeed

Figure 2.3: Classical Cassegrain antenna schematic.

As a first approximation, exploration and target tracking will be performed using amechanical approach that re-steers the main beam by rotating the entire antenna in azimuth andelevation, according to the monopulse error signals. In the future, different possibilities will beevaluated in order to substitute this mechanical scanning by an electronic exploration scheme.

8 2.3. SPADERADAR ARCHITECTURE

Chapter 5 includes the characterization of a reflectarray that could substitute the hyperbolicsubreflector of the Cassegrain system to provide electronic beamsteering in two dimensions [11].There is still much work to be done since the reflectarray sample that has been characterized isthe just first prototype of the final antenna that could be employed in the radar.

Chapter 3

Antenna System Design

This chapter describes the design of an antenna system at 94GHz for the space debris radar.Considering the radar specifications, a Cassegrain antenna system has been optimized for theapplication.

Different antenna geometries are analytically evaluated in Section 3.3 according to traditionaldesign criteria [12]. Then, those designs are numerically evaluated in Section 3.4 using thesimulation tool Grasp (General Reflector and Antenna Farm Analysis Software R©). At the endof the chapter, in Section 3.5, the final design for the Cassegrain antenna system is presented.

3.1 The Spaderadar Antenna System

This application requires an antenna capable of producing an extremely narrow beam withmore than 55 dB gain at 94GHz. Besides, the system will be monopulse to gain target trackingprecision. Table 3.1 presents a summary of the antenna system specifications.

Parameter Value

Operating frequency 94GHz

Bandwidth > 2GHz

Gain > 55 dB

Polarization RHCP

Weight/Size Minimum

Exploration Mechanical

Monopulse Tracking Yes

Table 3.1: General specifications of the radar antenna system.

The necessary antenna gain to maintain a moderate transmitted power and still cover morethan 10 km range was determined in [2]. Besides, since this is a space-borne application, it isconsiderably important that the entire antenna system is as compact and light in weight aspossible.

9

10 3.1. THE SPADERADAR ANTENNA SYSTEM

3.1.1 Cassegrain reflector system

Double reflector antennas are utilized for different applications, since they retain someadvantages with respect to ordinary single-reflectors [13]. For this specific radar application,an axially symmetric Cassegrain antenna configuration has been selected due to its compactnessand the possibility of placing the feed and the transceiver behind the main reflector.

A schematic of the classical Cassegrain configuration is presented in Figure 3.1. The feedwould be directly connected to the transmitter/receiver and pointed at a hyperbolic subreflectorwhich is suspended in front of a larger main parabolic reflector. This antenna is designed toachieve a uniform phase front in the aperture of the paraboloid and it can obtain efficiencies of65 - 80% (if surfaces are shaped [14]).

Figure 3.1: Centred Cassegrain antenna geometry [12].

The analysis of this antenna system is simple and can be described by only four independentparameters [10]:

• The main reflector profile depends on the parameter F , its focal distance. In [12], itssurface is defined by Equation 3.1.

• The subreflector is characterized by its eccentricity e and its focal distance f . Alternatively,Granet [12] uses the semi-transverse axis a of the hyperbola instead of the eccentricity.The surface of the subreflector is defined by Equation 3.2.

z + F =x2 + y2

4Fwith: x2 + y2 ≤ D2

4(3.1)

z + f = a

√1 +

x2 + y2

f2 − a2with: x2 + y2 ≤ d2S

4and e =

f

a> 1 (3.2)

Through geometrical optics, it can be shown that this arrangement folds the rays so that thewaves that emanate from the phase center of the feed illuminate the subreflector and are reflectedby it towards the other focus of the hyperboloid. Since this other focus of the hyperboloid isco-located with the focus of the paraboloid, rays propagate towards the primary reflector as if

CHAPTER 3. ANTENNA SYSTEM DESIGN 11

they were originated at the focal point of the paraboloid. Therefore, these rays are reflected bythe primary reflector and transformed into parallel rays [10].

Any dual reflector system can be considered as being replaced by an equivalent singlefocusing surface, which in the case of a Cassegrain system is another paraboloid with larger focallength (since the hyperboloid slows the beam divergence). The focal distance of the equivalentsingle paraboloid Fe depends on the eccentricity of the subreflector e through the magnificationparameter M [15].

M =e+ 1

e− 1(3.3)

Fe = M · F (3.4)

3.1.2 Monopulse feed

Different proposals for the monopulse feed of the Cassegrain system are already being designedby the GEA (Applied Electromagnetism Group). It will consist of a multiflare horn attachedto a mode extractor that will produce the sum and difference responses that are required ina monopulse system. These designs are not part of the objectives of this Thesis, though it isimportant to know that first pre-designs are producing return losses of 20 - 25 dB and isolationbetween ports above 30 dB [16].

3.2 Design Criteria

The gain of a Cassegrain antenna, in which the shadow of the subreflector over the main dish islarger than the feed aperture, is given by Equation 3.5. Overall antenna efficiency η is usually65 - 80% and it includes different effects such as illumination efficiency, spillover losses, losses inthe conductors, diffraction losses, blockage by the struts, etc.

G =4π

λ2Aeff = η

π2(D2 − d2S

)λ2

(3.5)

The following list summarizes some design criteria that maximize the overall efficiency of aCassegrain antenna for a certain application [12, 17]:

• Main dish should have at least 50λ diameter and F/D ratios between 0.25 and 0.8.

• The subreflector diameter dS should be larger than 5λ in order to avoid excessive diffractionlosses.

• The subreflector diameter dS should be smaller than 20% of the size of the main reflectordiameter, for the purpose of obtaining high blockage efficiency and avoiding excessivelyhigh sidelobe levels.

• The subreflector must be in the farfield of the feed (f + a > 2d2S/λ). Otherwise, there willbe significant phase errors.

• The semi-subtended angle ΨS with which the feed illuminates the subreflector should bechosen to obtain an edge illumination at the subreflector in the range of −10 to �12 dB.

12 3.2. DESIGN CRITERIA

3.2.1 Restricting dimensions

Section 3.1 establishes some specifications on the antenna system that is being developed.Analyzing this specifications from the point of view of a Cassegrain geometry produces somerestrictions on certain dimensions of the design [2].

Gain −→ Main diameter

Figure 3.2 has been obtained using Equation 3.5 to obtain an estimation of the Cassegrainantenna gain. Since the specification for the gain involved having at least 55 dB, a main reflectordiameter of 900mm has been chosen as fixed dimension for the pre-designs. This value leavessome margin of about 2 dB to face possible reduction in the performance.

400 600 800 1000 1200 1400

D (mm)

50

52

54

56

58

60

62

G (d

B)

Gain vs Diameter

Eff=0.8, ds=10%Eff=0.7, ds=10%Eff=0.6, ds=10%Eff=0.8, ds=20%Eff=0.7, ds=20%Eff=0.6, ds=20%

Figure 3.2: Gain curves with respect to the main reflector diameter.

Feed position

The dimensions for the future horn can be approximated by the schematic from Figure 3.3. Asobserved, the feed must be close to the apex of the main paraboloid so that the monopulsecomparator and the transceiver lie behind the main reflector. For the following pre-designs thefeed phase center has been assumed to be located 20mm in front of the main reflector.

28-30 mm

30-40 mm

20-30 mm

24-26 mm

Figure 3.3: Position and approximate dimensions of the monopulse horn.


Feed directivity −→ Semi-subtended angle

Currently there are two possible pre-designs of the feed horn that are being developed in paralleland their performance will be compared in the following pre-designs.

• Feed with 21.5 dB directivity in the sum pattern.

• Feed with 24.8 dB directivity in the sum pattern.

Approximating the radiation patterns of those feeds as an ideal cosq model [10], it can beshown that the semi-subtended angle ΨS to obtain �10 dB taper is 14.6◦ for the horn with21.5 dB directivity and 10◦ for the horn with 24.8 dB directivity. The derivation is presented inEquation 3.6 and the resulting directivity in Figure 3.4.

E(Φ, θ) = cosq(θ) with: D0 =4π

Ω= 2(2q + 1) −→ q =

D0 − 2

4(3.6)

-30 -20 -10 0 10 20 30

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-40

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-25

-20

-15

-10

-5

0

E (d

B)

Feed directivity model

21.5 dB (q = 34.81)24.8 dB (q = 75)

Figure 3.4: Feed directivity model.

Compactness and cost −→ Focal length

Since this is a spaceborne application, it is important that the final system is as compactand light in weight as possible. The total length of a Cassegrain system is calculated withEquation 3.7 [12].

Ltot = F + a

√1 +

d2S4(f2 − a2)

− f (3.7)

Total length is highly dependent on the focal distance of the main reflector, so lower F/Dratios are expected to produce more compact systems. Indeed usual F/D ratios for Cassegrainantennas range between 0.3 and 0.5, although for satellite and monopulse applications shallowerdishes are used with F/D up to 1. The reason lies on the fact that lower profile paraboloids areeasier to support and to move mechanically, since they require less material for their fabrication,which also makes them lower-prized [18].

The following sections will compare the performance of F/D ratios of 0.4 and 0.75 in orderto decide which range of F/D is best for this application.

14 3.3. ANALYTICAL SOLUTION

3.2.2 Case studies

According to the antenna system restrictions detailed along this section, Cassegrain optimumdesigns for two different feed directivities and two different F/D ratios will be compared inSection 3.3 and Section 3.4. Feed location and main dish diameter have been set to reasonablevalues given the system specifications. The four different combinations that will be analyzedalong following sections are presented in Table 3.2.

CaseMain dishdiameter

Feedposition

Edge taperillumination

F/D ratioFeed

directivity

1

900mm 20mm �10 dB

0.4 21.5 dB

2 0.4 24.8 dB

3 0.75 21.5 dB

4 0.75 24.8 dB

Table 3.2: Study cases for the Cassegrain antenna system.

3.3 Analytical Solution

The purpose of this section is to determine the geometrical dimensions and the expectedperformance of the four cases proposed in Table 3.2 using analytical expressions.

3.3.1 Analytical equations

According to Section 3.2, for this application there are four parameters that restrict the systemgeometry : D, F , Lm and ΨS (see Figure 3.1). For those four input parameters, Granet [12]provides a set of equations to find the rest of the variables that define the geometry of thesystem:

a =1

2

(Lm − F )

[−D + 4F tan

(ΨS

2

)]

D + 4F tan

(ΨS

2

) (3.8)

Ls = − D(Lm − F )

D + 4F tan

(ΨS

2

) (3.9)

f =1

2(F − Lm) (3.10)

dS =

16 sin(ΨS)DF (Lm − F )

[−D + 4F tan

(ΨS

2

)]

8FD

[D + 4F tan

(ΨS

2

)]+ (D2 + 16F 2) sin(ΨS)

[D + 4F tan

(ΨS

2

)] (3.11)


Using Granet’s Equations 3.8, 3.9, 3.10 and 3.11 optimal Cassegrain geometries have beenfound for the four cases from Table 3.2. The obtained results are presented in Table 3.3.

Case 1 Case 2 Case 3 Case 4

Input parameters

Main dish diameter, D 900mm 900mm 900mm 900mm

Main dish focal distance, F 360mm 360mm 675mm 675mm

Directivity, D0 21.5 dB 24.8 dB 21.5 dB 24.8 dB

Semi-subtended angle ΨS at C = −10 dB 14.65◦ 10.01◦ 14.65◦ 10.01◦

Feed position, Lm 20mm 20mm 20mm 20mm

Output parameters

Subreflector diameter, dS (3.11) 157.7mm 110.5mm 253.9mm 187.2mm

Subreflector semi-transverse axis, a (3.8) 112mm 128.2mm 145.2mm 191.2mm

Subreflector focal distance, f (3.10) 170mm 170mm 327.5mm 327.5mm

Eccentricity, e (3.2) 1.52 1.33 2.26 1.71

Distance from feed to subreflector, Ls (3.9) 282mm 298.2mm 472.7mm 517.7mm

Total length of the antenna, Ltot (3.7) 302.2mm 317.8mm 492.8mm 535.8mm

Estimated gain (3.5) 57.84 dB 57.91 dB 57.62 dB 57.79 dB

Equivalent focal distance, Fe (3.4) 1750mm 2569mm 1750mm 2569mm

Magnification, M (3.3) 4.86 7.14 2.59 3.81

Table 3.3: Analytical results for different specifications.

3.3.2 Analysis of the results

• A Cassegrain system using an F/D = 0.4 results in an antenna system much more compactthan the shallower system with F/D = 0.75. However, lower profile paraboloids will belighter in weight so one advantage may compensate the other.

• Subreflector diameters are sufficiently large in terms of wavelengths (dS > 5λ) so thatdiffraction losses are below 0.2 dB.

• When the feed position is fixed, large focal lengths or low feed directivities produce largersubreflector diameters. Therefore, those designs are producing higher blockage, whichexplains the small loss in performance observed in the estimated gain.

• Estimated antenna gain is very similar for all the designs, obtaining a maximum of 57.9 dBfor the case of F/D = 0.4 with a 24.8 dB horn. The utilized overall efficiency is a coarseestimation of spillover, illumination and blockage efficiency.

The estimated performance of these designs has been evaluated in terms of the estimatedgain. This figure of merit is neither sufficient, nor accurate for making a decision, so furtheranalysis is required by using a numerical simulator.

16 3.4. NUMERICAL SOLUTION

3.4 Numerical Solution

Since reflector antennas usually have 10 - 1000λ, the sub-wavelength gridding that is required infull-wave solutions would produce extremelly large problems. In order to reduce computationalcomplexity, it is necessary to resort to approximate methods.

This section utilizes the simulation tool Grasp to characterize more accurately theperformance for the different cases proposed in Table 3.2. Grasp is a trademark reflector analysistool from TICRA that is able to predict the electromagnetic fields produced by high frequencyreflector systems. This is attained by solving the time-variant Maxwell’s equations using highlyefficient numerical approximations known as PO (Physical Optics) and PTD (Physical Theoryof Diffraction) [19]. Appendix A summarizes the behaviour of those algorithms when they areused to simulate reflector antennas.

3.4.1 Simulation set-up

Appendix A details how to configure Grasp to simulate the farfield of a Cassegrain system.

• Simulations that are performed within this section include:

– Spillover efficiency.

– Illumination efficiency.

– Subreflector blockage efficiency.

– Diffraction at the reflector edges.

• Simulations that are performed within this section do not include:

– Blockage and diffraction caused by the supporting structures (they will be very similarfor the four cases).

– Losses and roughness of the conductor surfaces.

– Other possible non-idealities.

The ideal Gaussian feed model parameters to simulate each case are described in Table 3.2are presented in Table 3.4.

Case Taper angleTaper

illumination Polarization

1 14.6◦

�10 dB RHCP2 10◦

3 14.6◦

4 10◦

Table 3.4: Gaussian feed configuration for each of the four comparedgeometries so that illumination at the edge of the subreflector is �10 dB.

The analyzed theta range from �180◦ to 180◦ is configured to use 10001 points. Since boththe feed and the reflector system have circular symmetry, all the cuts of the radiation patternin phi are equal.


3.4.2 Simulation results of different pre-designs

Geometrical dimensions determined in Table 3.3 for the four pre-design cases are introduced inGrasp and simulated. The distinct geometries are illustrated in Figure 3.5, where each pre-designis fed as indicated in Table 3.4.

(a) (b) (c) (d)

Figure 3.5: Analyzed Cassegrain geometries from Table 3.2: (a) case 1,(b) case 2, (c) case 3 and (d) case 4.

Results obtained after having simulated each of the four proposed cases are summarized inTable 3.5. The resulting radiation patterns are respectively illustrated in:

• Figure 3.6 for case 1 −→ F/D = 0.4 and 21.5 dB-feed.




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Farfield versus frequency

Copolar 92 GHzCrosspolar 92 GHzCopolar 94 GHzCrosspolar 94 GHzCopolar 96 GHzCrosspolar 96 GHz

(c)

Figure 3.6: Simulated farfield results for geometrical design 1: (a)Copolar, (b) crosspolar and (c) zoomed response.


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Parameter Case 1 Case 2 Case 3 Case 4

Focal distance, F/D 0.4 0.4 0.75 0.75

Feed directivity 21.5 dB 24.8 dB 21.5 dB 24.8 dB

Gain 57.4 dB 57.56 dB 56.59 dB 57.22 dB

Side Lobe Level �19.18 dB �20.96 dB �14.8 dB �17.85 dB

Crosspolar ratio �58.01 dB �55.61 dB �60.8 dB �60.6 dB

Beamwidth 0.231◦ 0.235◦ 0.221◦ 0.226◦

Subreflector diameter, dS/D 17.5% 12.3% 28.2% 20.8%

Total length of the antenna 302.2mm 317.8mm 492.8mm 535.8mm

Overall efficiency 70% 72.6% 58.1% 67.2%

Table 3.5: Comparison of the results obtained for different pre-designs.

Overall efficiencies have been calculated by comparing the simulated gain to the directivityof a paraboloid the same size as the main reflector. Values larger than 70% are very gooddesigns. Some times the reflectors can be shaped to obtain up to 80%.


3.4.3 Analysis of the results

Main results obtained within this section are summarized in Table 3.5.

• A main reflector with F/D = 0.4 produces better results, in terms of gain and secondarylobes, than a main reflector with F/D = 0.75.

• All the analyzed cases obtain simulated gains larger than the specification. As observedin the radiation patterns, neither is the bandwidth restriction an issue.

• Obtained crosspolar ratios are quite low for all the designs, although they are better forthe lower profile antennas of F/D = 0.75.

• As it was obtained within Section 3.3, the overall length Ltot of the antenna system issmaller for F/D = 0.4. Besides, lower directivity feeds do also produce more compactsystems, since the beam has to be interrupted sooner to prevent it from diverging toomuch.

• Since it has been required that the feed is considerably close to the apex of the mainreflector, for F/D = 0.75, imposing this criterion and obtaining the required subtendedangle generates a subreflector diameter that is considerably large. This produces higherblockage and, therefore, performance losses in terms of gain and sidelobe levels.

• For the same reason as the previous point, when the low directivity feed is used, thesubreflector diameter tends to grow to accommodate to the larger subtended angle that isrequired to maintain an optimal illumination of C=�10 dB.

As a summary, these simulations show that the best performance is expected to be obtainedfor a Cassegrain system using a main paraboloid of F/D = 0.4 illuminated by a feed with 24.8 dBdirectivity. Shallower paraboloids produce higher secondary lobes and blockage losses due totheir large subreflectors. The geometrical and electrical parameters for this best pre-design (case2) are collected in Table 3.6.


Parameter Value

Specifications

Main dish diameter, D 900mm

Main dish focal distance, F 360mm

Feed directivity, D0 24.8

Semi-subtended angle, ΨS 10.01◦

Feed position, Lm 20mm

Geometrical parameters

Subreflector diameter, dS 110.5mm

Subreflector semi-transverse axis, a 128.2mm

Subreflector focal distance, f 170mm

Eccentricity, e 1.33

Distance from feed to subreflector, Ls 298.2mm

Total length of the antenna, Ltot 317.8mm

Equivalent focal distance, Fe 2569mm

Magnification, M 7.14

Simulated results

Gain 57.56 dB

Side Lobe Level �20.96 dB

Crosspolar ratio �55.61 dB

Beamwidth 0.235◦

Efficiency 72.6%

Table 3.6: Geometrical and simulated results of the chosen configuration.

3.5 Final Antenna Design

After having encountered the range of focal distances and feed directivies for which the antennasystem works the best, the final optimization has been performed by experts from the GR(Radiation Group) of UPM. Although the optimization process is not part of the work performedwithin this Thesis, this section presents the final antenna geometry for the space debris radarand the simulated results.

3.5.1 Geometrical definition

The geometrical parameters of the final reflector system are presented in Table 3.7. The resultinggeometry is shown at Figure 3.10.

Conformation strategies over the reflector and subreflector dishes were also considered forobtaining a more uniform illumination while maintaining low spillover losses, thus increasing theoverall efficiency. However, the application of conformation has been finally dismissed, becauseit would increase the secondary side lobes.

22 3.5. FINAL ANTENNA DESIGN

Parameter Value

Main dish diameter, D 900mm

Main dish focal distance, F/D 0.36

Main dish focal distance, F 324mm

Semi-subtended angle, ΨS 9.35◦

Feed position, Lm 34mm

Subreflector diameter dS/D 10%

Subreflector diameter, dS 90mm

Subreflector semi-transverse axis, a 114mm

Subreflector focal distance, f 145mm

Eccentricity, e 1.27

Distance from feed to subreflector, Ls 259mm

Total length of the antenna, Ltot 293mm

Table 3.7: Geometrical parameters of the final Cassegrain system.

Figure 3.10: Geometry of the final design of the dual antenna system.

3.5.2 Simulation results

The proposed geometry has been simulated in Grasp utilizing the configuration described inAppendix A. The simulation results are presented in Table 3.8 and Figure 3.11.

Parameter Value

Gain 57.5 dB

Side Lobe Level �20.71 dB

Crosspolar ratio �52.67 dB

Beamwidth 0.235◦

Efficiency 71.6%

Table 3.8: Simulation results for the final Cassegrain design optimallyilluminated by a Gaussian feed.


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Figure 3.11: Simulated farfield of the final design illuminated by C =−10 dB: (a) Copolar (RHCP), (b) crosspolar (LHCP) and (c) zoomedfrequency response.

3.5.3 Considerations on the final antenna system

This design was optimized by the Radiation Group taking into account further aspects of thedesign than those observed in Table 3.8. For instance, ease of fabrication, supporting structures,conductor losses or blockage by the struts were considered. Therefore, the simulated resultsshould not be compared with those obtained in previous sections, since they are based on morecomplex considerations. Besides, the design was optimized for a third model of feed horn with adirectivity of 24.55 dB, to which a correcting lens might be attached to improve overall systemefficiency.

As a task for the future, the fields produced by the monopulse feed that is being designedwill be validated for this reflector system. For this purpose, a Grasp object called “tabulatedfeed” can be used to introduce the real sum and difference patterns of the monopulse horn as afeed of the Cassegrain antenna.

24 3.5. FINAL ANTENNA DESIGN

Chapter 4

Software Tool Design for SimulatingQuasi-optical Measurement Systems

The simplest free-space measurement system consists of a pair of horns and a sample locatedbetween them. However, in most optical applications it is necessary to focus, modify, or shape thebeam by using lenses, mirrors and other optical elements. For this reason, this chapter developsa Matlab software tool based on Gaussian beam propagation theory that can be utilized for theanalysis and synthesis of quasi-optical test bench designs.

In order to examine the capabilities of the developed tool, Section 4.3 offers different examplesof reflection-based measurement test benches that could be utilized to measure the reflectioncoefficient of a sample at W-band.

4.1 Theoretical Background on Quasi-Optical Systems

Quasi-optical systems are a suitable alternative to guided transmission lines at sub-millimeterwavelengths, since not only do they provide exceptionally low losses, but they can also handlemultiple polarizations and large bandwidths.

Quasi-optical system design is based on Gaussian beam propagation theory. By means ofre-focalizing the beam using lenses or mirrors, Gaussian propagation can distribute the inputpower through a region of several wavelengths [20].

4.1.1 Gaussian beam propagation in free space

Considering that diffraction is small with respect to a wavelength and that axial variations of thefield are negligible, the wave equation is reduced to what is called the paraxial wave equation.This approximation can be applied when the angular divergence of the beam is reasonablyconfined to 30◦. The solution of the paraxial equation are the Gaussian beam modes, whichare used as the basis of quasi-optical design [21]. The fundamental and most simple transverseGaussian mode has the normalized Gaussian field presented in Equation 4.1.

E(r, z) =

√2

πw2(z)exp

(−jkz − jπr2

λR(z)− r2

w2(z)

)(4.1)

The first exponential factor of Equation 4.1 describes the phase of a plane wave, the second

25

26 4.1. THEORETICAL BACKGROUND ON QUASI-OPTICAL SYSTEMS

factor is responsible for the phase front curvature and the last exponential factor determines thefield intensity in the transverse direction [22].

There are some basic parameters that describe a Gaussian beam according to this model:

w(z): The radius w of a Gaussian beam is defined as the lateral distance from the propagationaxis of the beam where the field amplitude has decayed by exp(−1). The “beamwaist” w0

is the minimum beam radius of the beam.

R(z): The radius of curvature R is defined as the radius of curvature of the spherical wavefrontsthat describe the equiphase surfaces of the different modes.

θ0: The variation of the beam radius w as a function of distance z from the beamwaist hasthe form of a hyperbola. The asymptotic growth angle of the beam radius is θ0.

zc: Rayleigh distance zc is the distance relative to the beamwaist position, in which thebeam can be said to be approximately collimated. It is a useful quantity to approximatethe beam parameters in two different regions: z << zc is the near-field region, whilez >> zc is the farfield region.

The parameter z defines the propagation axis. As the beam propagates towards thebeamwaist position, the beam is said to be converging, and after the beamwaist, the beam isdiverging. At the beamwaist position, the beam radius is minimum and the radius of curvatureis ±∞ (see Figure 4.1).

zc

Beamwaist

w0

Beam

diverges

Beam

converges

Plane

wavefront

R→∞ Spherical

wavefront

Spherical

wavefront

θ0-zc

Rmin = 2 zcRmax = -2 zc

w = √2 w0w = √2 w0

Figure 4.1: Evolution of the beam radius (black) and phase radiusof curvature (green) for a Gaussian beam that propagates in a certaindirection.

The variation of beam parameters with distance z, measured from the beamwaist, is givenby the following equations[21]:

w(z) = w20 ·

(1 +

(z

zc

)2)

(4.2)

R(z) = z +z2cz

(4.3)

tan(θ0) =z

zc(4.4)

zc =πw2

0

λ(4.5)

CHAPTER 4. SIMULATION OF QUASI-OPTICAL MEASUREMENT SYSTEMS 27

4.1.2 Gaussian beam transformation

The treatment of rays in linear optical geometrical systems is based on ABCD transformationmatrices, which are defined for each optical element. The quantities of interest are the positiondin and the slope d′in of the ray in the input plane, which are transformed into the correspondingquantities dout and d′out of the ray in the output plane . Since the radius of curvature is definedas R = position/slope, the relationship from Equation 4.7 can be defined [23].

[dout

d′out

]=

[A B

C D

]·[din

d′in

](4.6)

Rout =doutd′out

=ARin +B

CRin +D(4.7)

The extension of this ray transformation approach to Gaussian beams, leads to an ABCDlaw in which the same matrices operate on the complex radius of curvature q which is definedin Equation 4.8 [24]. The complex radii of curvature of the beam at the input and the outputplanes of an optical structure are related by the transformation of Equation 4.9 [23].

1

q(z)=

1

R(z)− i

λ

πw2(z)=

1

z + i zc(4.8)

qout =Aqin +B

Cqin +D(4.9)

The equivalent matrix for a sequence of elements can be obtained by multiplying theindividual ABCD matrices, starting with that for the first element encountered by the beamand adding the matrix of each subsequent element on the left [21].

ABCDtotal = ABCDn ·ABCDn−1 · (· · · ) ·ABCD2 ·ABCD1 (4.10)

Gaussian Beam Sources

Waveguide horns are among the most common radiating elements that produce an approximatelyGaussian radiation pattern. A complete list of the approximate beam radii at the aperture ofvarious feed structures is presented by Goldsmith in [20]. This can be used when accuratemeasurements of the beam size are not available.

Knowing the beam radius at the aperture (w(zap)) and assuming that the radius of curvatureat the aperture (R(zap)) is equal to the slant length of the horn, the beamwaist radius w0 andits position behind the aperture z0 can be approximated using Equation 4.11 and Equation 4.12[21].

w0 =w(zap)√

1 +πw2(zap)

λR2(zap)

(4.11)

z0 =πw0

λ

√w2(zap)− w2

0 (4.12)

28 4.1. THEORETICAL BACKGROUND ON QUASI-OPTICAL SYSTEMS

Beam transformation by a lens

The ABCD matrix of a thin lens, whose surfaces have radii of curvature R1 and R2, is definedusing Equation 4.13 [24].

ABCDlens =

⎡⎣ 1 0

− 1

F1

⎤⎦ with:

1

F=

nlens − n0

n0

(1

R2− 1

R1

)(4.13)

When this matrix is applied to Equation 4.9, the specific transformation produced on aGaussian beam after going through a thin lens is found to be modelled by Equations 4.14 and4.15.

dout = F ·

⎡⎢⎢⎢⎢⎣1 +

dinF

− 1(dinF

− 1

)2

+

(zcF

)2

⎤⎥⎥⎥⎥⎦ (4.14)

w0 out =w0 in√(

dinF

− 1

)2

+

(zcF

)2(4.15)

A thick lens (it has a certain thickness d) can use those thin lens equations if a certainequivalent focal distance is defined as indicated in Figure 4.2b. In practice, it is also importantto know the front and back focal lengths of the lens, Ffv and Fbv [25].

din dout

w0outw0in

(a)

Fe Fe

Ffv Fbv

h2h1

d

(b)

Figure 4.2: Gaussian beam transformation by lens: (a) thin lens modeland (b) modified model for a thick lens.

Beam transformation by a focusing mirror

In quasi-optical systems, mirrors provide an alternative to lenses as focusing elements for beamradiation. The beam transformation that occurs at the surface of a mirror is analogous to whathappens in a thin lens, excepting for the folding of the propagation axis. Therefore, Equations4.14 and 4.15 can be used to determine the reflected beam parameters of a mirror [23].

• A spherical mirror with a curvature radius R would have the ABCD matrix of

Equation 4.13, with1

F=

2

R.


• An ellipsoidal mirror would have the ABCD matrix of Equation 4.13, with1

F=

1

D1+

1

D2,

being D1 and D2 the distances from each focus of the ellipsoid to the center of the mirrorsection.

• A paraboloid mirror is equivalent to an ellipsoid mirror that has a focus at the infinity.Therefore, it has the ABCD matrix of an ellipsoid particularized for D2 = ∞ and D1 =Fparab/cos(θ), where θ is the off-axis angle of the mirror.

Mirrors usually have to be off-axis resulting in the beam suffering from cross-polarization anddistortion effects, specially when the effective focal distance is small. These effects are quantifiedin [26] and will not be taken into account in the simulation tool.

Beam transformation by a telescope: the afocal system

Usually, when trying to take measurements over a large bandwidth, a non-frequency dependentoptical configuration is desired. In general, the relationship between the input beamwaist radiuswin and the output beamwaist radius wout depends on their locations din and dout, and theycan be derived using the ray transfer matrices of Equation 4.9. A special case occurs when theinput beamwaist position din is equal to the focal length F of the lens: Equation 4.14 showsthat the output beamwaist position dout will be equal to this focal distance F , regardless ofthe frequency. This fact is the basis of a “telescope transformation”, in which the beamwaistradius and its position can be controlled by separating every two lenses by the sum of their focallengths. This allows the creation of what is called an “afocal system”, since an object separatedan infinite distance from the first lens will form an image at infinity [25].

A “beam-waveguide” is an afocal system that takes advantage of a telescope-basedtransformation to guide waves as waveguides do. It injects a Gaussian beam at the focal pointof the first lens, so that the output beamwaist is located exactly at the output focal point ofthe last lens. If the output horn was located there, maximum power coupling from one hornto the other would be obtained. An example of an afocal quasi-optical system is presented inFigure 4.3. The sample (in green) that needs to be characterized is placed at the internal focalpoints of the lenses (they are coincident). In this way, it is guaranteed that a beamwaist orfocusing point will occur exactly at the position of the sample, which means that it would beilluminated by a plane wave maximally collimated regardless of the frequency of operation [25].

F1F1 F2

w0

w0

Figure 4.3: Example of an afocal system for quasi-optical beampropagation.

30 4.2. DEVELOPED GAUSSIAN BEAM TRACING TOOL

4.2 Developed Gaussian Beam Tracing Tool

A software tool based on Gaussian beam propagation theory has been developed using Matlabas a programming language. It propagates the beam in a two dimensional bench that can befreely configured using horns, lenses and mirrors as optical elements. This section explains howto correctly execute a simulation (Section 4.2.1), describes its operational flow (Section 4.2.2)and details the simulation outputs (Section 4.2.3).

4.2.1 Running the software

The following enumeration summarizes the steps to make a proper use of the implementedsoftware.

1. Open file MAIN.m.

2. Type frequency of operation in GHz.

3. Choose the exponent of the exponential field decay that will be plotted in addition to theexponent decay -1 (it corresponds to �8.7 dB power decay).

4. Choose which optical system components to use from those available.

5. Choose an arrangement of the optical elements from those that are pre-defined in the tool.

6. Choose separation between horns and lenses.

7. Choose separation between mirrors and the sample.

8. Execute file MAIN.m.

9. Wait for the results (see Section 4.2.3).

Some pre-designed configurations are already programmed and can be modified to testdifferent situations. If any different optical configuration is needed or any new componenthas to be simulated, some files must be modified or created.

4.2.2 Step-by-step simulation process

This section summarizes the basic operations that are sequentially performed by the simulatorduring execution time.

1. The initial parameters are entered as indicated in Section 4.2.1.

2. The corresponding optical system file is loaded:

2.1. Characteristic parameters of each utilized optical element are obtained.

2.2. Each optical element is positioned and rotated in a two-dimensional bench accordingto the configuration file.

3. Propagation axis is unfolded and beam parameters are iteratively calculated as the beampropagates between optical elements:


3.1. Relative input beamwaist position din is obtained.

3.2. Beam parameters, din and w0 in, are transformed using the corresponding the ABCDlaw (Equation 4.9) into the beam parameters referred to the output plane of thestructure, dout and w out.

3.3. Paraxial approximation is checked (w0 >> 0.9λ)

3.4. Absolute output beamwaist position is calculated from the relative distance dout.

3.5. Go back to step 3.1.

4. Rays and optical elements are plotted in the unfolded one-dimensional test bench. Thisapproximation makes mirrors behave as thin lenses.

5. Optical axis is re-folded according to the configuration file and rays and optical elementsare plotted in a schematic of a two-dimensional test bench.

6. Overall results of the simulation are calculated, plotted and saved in the correspondingfiles (see Section 4.2.3).

4.2.3 Simulation results and output files

When the simulation stops, results are stored into different files:

Figure 1: two-dimensional propagating beam. This figure shows a scaled version of theoptical bench in which the optical elements and the propagating beam are seen fromabove. The beam propagates from one element to the next. This figure is saved asResults/fig1.png and Results/fig1.fig.

Figure 2: simplified one-dimensional propagating beam. This figure unfolds the opticalaxis so that the beam propagates in a constant direction. This unfolding basically assumesnormal incidence over each surface and makes mirrors behave as lenses. This figure is savedas Results/fig2.png and Results/fig2.fig.

Output text file. This text file includes all the important results generated during thesimulation. These results include:

– Positions of the optical structures in the two-dimensional optical test bench.

– Amplitude taper of the beam at the edges of each optical structure.

– Beam parameters at the sample position.

– Estimated fraction of power than impinges at the sample and at the output horn (itonly considers spillover losses).

This file is saved as Results/output.txt.

4.2.4 Limitations of the simulation tool

The simulation tool that has been implemented along this chapter allows to predict theapproximate behaviour of a fundamental Gaussian mode propagating through an optical bench.The tool has many limitations, so it is not intended to obtain a detailed and accurate responseof the optical arrangements:

32 4.3. APPLICATION EXAMPLE: 45◦ INCIDENCE

• Beam transformations are calculated assuming normal incidence and that distancesbetween elements are given by the distances between their centers.

• For the calculation of the beam transformations along the optical axis, elements areassumed to be ideal: lossless, reflectionless and with an infinite diameter.

• Cross-polarization and distortions caused by the feed and the optical structures are notbeing taken into account.

• The pre-configured systems are perfectly aligned systems, since beams propagate alongthe unfolded optical axis of the system.

• Horns are not ideal Gaussian beam radiators. Modifications are required if higher orderGaussian modes need to be taken into account.

As a summary, this tool implements a very simplified version of the Gaussian beampropagation theory. Modifications based on 3 × 3 or 4 × 4 ABCD matrices are not difficultto incorporate in order to produce more accurate results that take into account misalignments,losses, reflections or crosspolarization [27]. Other approximations of Gaussian beam theory usea mode matching method to calculate the transmitted and reflected Gaussian modes at eachoptical structure [23].

4.3 Application Example: Reflection-based Optical Set-ups toMeasure Samples at 45◦ Incidence

In order to examine the capabilities of the developed tool, this section proposes different opticalconfigurations that could be used to measure the reflection coefficient of a sample impinged atan angle of 45◦ from boresight.

Each configuration produces a different propagation pattern, ones being more collimatedat the sample position and others less collimated. Therefore, each configuration willrequire a different minimum sample size to maintain a certain constraint on edge diffractionand transferred power. For the purpose of making this analysis extensible to any futuremeasurements, the main goal of this analytical example has consisted of obtaining the rangeof sample sizes for which each quasi-optical bench could produce accurate measurements.

4.3.1 Design criteria

The criteria for the test bench designs of this section are summarized in the following list:

• In order to guarantee good results across different frequencies, these set-ups have beendesigned as afocal systems, whose characteristics were explained in Section 4.1.2.

• Illumination at the edge of the optical elements should be as low as possible, so that mostenergy is transmitted to the next element and border diffraction effects are minimized. Anusual requirement for this taper illumination is �17.4 dB at the border, which is equivalentto a exp(−2) field decay [21].

• Physical elements utilized in the optical bench must not overlap.

• Optical elements should interact only once with the propagating beam.

• The propagating beam should not intersect with itself.


4.3.2 Simulated optical configurations

Considering the design criteria from Section 4.3.1, six different optical configurations areproposed for measuring the reflection coefficient of a flat sample impinged at an angle of 45◦

from boresight:

• Using two horns and two dielectric lenses: Section B.1.

• Using two horns and two 45◦ off-axis mirrors: Section B.2.


• Using two horns, two lenses and two 45◦ off-axis mirrors: Section B.4.

• Using two horns, two 45◦ off-axis mirrors and two 90◦ off-axis mirrors: Section B.5

– Option 1: Section B.5.1.


These configurations have been simulated using the Matlab software described in Section 4.2,specifically at 94GHz and 100GHz, which are the frequencies at which the Spaderadar Projectcomponents usually operate. Besides, two different models of horns are compared: a pair of23 dB RPG Potter horns and a pair of 21 dB Millitech standard conical horns. The selectedoptical elements are all available at the laboratory and their datasheets can be found inAppendix G. The detailed results of all the performed simulations belonging to each of theproposed configurations are presented in Appendix B.

4.3.3 Critical analysis of the simulation results

As an example of an scenario in which the developed Gaussian beam tracing tool could beutilized, different optical test benches have been designed to measure the reflection coefficientof a flat sample impinged from 45◦ with respect to its normal. The simulation results arepresented in Appendix B and their analysis will be useful in Chapter 5 to choose the bestfree-space measurement system to characterize a reflectarray antenna.

In all the designed set-ups the sample is located at a beamwaist position regardless ofthe frequency, which is a positive feature in case of a broadband measurement. Besides, theusual criteria to consider that the Gaussian beam approximation is sufficiently accurate for theapplication demands that the amplitude taper at the edges of every optical element is below�17.4 dB. If this is fulfilled, beam truncation effects are negligible [20]. Because of this theobjective was to calculate the sample size that obtains exactly those �17.4 dB at its border.

The following list summarizes some general conclusions that can be obtained from theanalyzed examples:

• Millitech conical horns produce a smaller beamwaist at their phase center than the RPGdual mode horns. This is because the aperture of the RPG horns is larger [21]. It hasbeen noticed that the beam radius at the sample position is smaller in case of:

– Using two focusing elements and the RPG horns.

– Using four focusing elements and the Millitech horns.

34 4.3. APPLICATION EXAMPLE: 45◦ INCIDENCE

• In all the analyzed designs the beam radius at the sample position is smaller at 100GHzthan at 94GHz. However, it has been noticed that, in a 5% bandwidth, actual differencesin the beam evolution throughout the bench are small.

• None of the optical configurations using 90◦ off-axis mirrors obtains less than �17.4 dBspillover losses at those mirrors. The reason for this is that the beam diverges too muchwhen it propagates along the distance from the horn too the mirror (equivalent focal lengthof the mirror), which could be avoided using horns with larger beamwaist at their phasecenters (e.g., with higher gain).

• Configurations using two 45◦ and two 90◦ off-axis mirrors do not comply with theparaxial approximation at the sample position, so the Gaussian beam approximation isnot representative of the real behaviour of the fields.

As a summary, the sample under test has to be larger than the minimum sample size toavoid truncation effects. Besides, if it is too large, only the response of the central part of thesample will be actually measured. A good trade-off between maintaining low truncation effectsand obtaining high illumination efficiency would demand from �17.4 dB to �25 dB at the edgeof the sample. Therefore, different configurations would be optimum for measuring differentsamples depending on their sizes. The sample sizes for the most promising configurations,according to simulations, are presented in Table 4.1.

ConfigurationSample size

94GHz 100GHz

Lenses (×2) 57 - 69mm 55 - 66mm

45◦ off-axis mirrors (×2) 108 - 130mm 104 - 125mm

Lenses (×2) + 45◦ off-axis mirrors (×2) 35 - 42mm 34 - 41mm

Table 4.1: Recommended sample sizes for using different free-spacemeasurement systems (using RPG horns).

Finally, another aspect to be considered in order to choose the best optical set-up is theavoidance of6 elements that introduce losses or uncontrolled distortions. For example, thelenses available at the laboratory introduce about 20% losses according to their datasheet (seeAppendix G).

Chapter 5

Characterization of a W-bandreflectarray

This chapter is aimed at the characterization of a reflectarray working at W-band. This antennawas designed by the Applied Electromagnetism Group and the Photonics Group of ETSIT(UPM) as a first prototype of a reflectarray to substitute the mechanical exploration scheme ofthe basic antenna system described in Chapter 3.

The software tool developed in Chapter 4 will be utilized to design the quasi-opticaltest benches necessary to measure the reflectarray sample. According to simulations, thoseconfigurations expected to obtain the best performance will be implemented to perform themeasurements (Section 5.3).

The measurements performed within this chapter are intended to serve as a general overviewof the behaviour of the reflectarray, in order to validate the designing and manufacturingmethods [28]. The main goal of the measurements presented in Section 5.4 has been to establishwhich excitations to apply to produce a certain quasi-permanent state at each unit cell ofthe reflectarray. Once this is obtained, pattern synthesis theory could be applied to obtainbeamforming and beamsteering capabilities.

5.1 Theoretical Background on Reflectarray Antennas

A reflectarray is an antenna consisting of a flat reflecting surface formed by many radiatingelements that are illuminated by a feed antenna (see Figure 5.1). When the feed illuminatesthe reflectarray elements, most of the energy is re-radiated. By modifying certain geometricalparameters of the radiating elements, the incident field at each elemental cell will be scatteredwith the electric phase required to form a planar wavefront in the farfield distance [29].

Each elemental cell must introduce a phase shift to compensate for the phase differencesassociated to the different paths between the horn and the different cells. Besides, it is possibleto introduce further phase shifts to apply array synthesis techniques and accurately tilt orconform the beam [30].

35

36 5.1. THEORETICAL BACKGROUND ON REFLECTARRAY ANTENNAS

Figure 5.1: Schematic of a reflectarray antenna [29].

5.1.1 Reflectarray antennas based on patches

The simplest elements that can be used in reflectarray antennas consist of variable-size printedpatches. By varying the resonant dimensions of the patches, the phase shift introduced by eachcell in the incident field can be modified [29].

A reflectarray made of a single layer of microstrip patches exhibits a narrowband behaviour.There are several techniques to increase the operational bandwidth:

• Increasing the substrate thickness (although this reduces the phase range).

• Stacking two or more array layers.

• Using more complex multi-resonant cells.

5.1.2 Reconfigurable reflectarrays based on liquid crystal

The principle of operation of a reconfigurable reflectarray is based on the possibility of varyingthe geometrical parameter that controls the phase shift (e.g., the resonant length) of the unit-cellelements.

LC (Liquid Crystal) is a thin membrane material that has the ability to change itspermittivity (dielectric anisotropy) when a quasi-static electric field is externally applied. Thisimportant feature is based on a property of nematic LC molecules to twist themselves to varyingdegrees depending on the applied voltage [31, 32].

If the LC is inserted in the gap between the printed patches layer and the ground layerof a reflectarray antenna, this tunable permittivity will modify the effective wavelength of thecell. Thus, LC properties can be exploited to transform each cell of the reflectarray antennainto an electronically controlled phase shifter [31, 33]. If each reflectarray cell can be biasedindependently with a certain voltage, the antenna could be used for 2D-scanning using arraysynthesis techniques [30].

Different strategies have already been successfully utilized to address the cells (pixels) of LCD(Liquid Crystal Display) screens [32]. They are usually based on time-multiplexing techniques,in which different cells are interconnected in rows at one side of the LC layer and in columns atthe other side, and then sequentially biased as required. These techniques could be adapted tooperate in reflectarray technology.

• Passive addressing: Due to the inherent capacitive behaviour of the LC, short pulses canbe applied periodically to each column and row to maintain the molecular twisted state

CHAPTER 5. CHARACTERIZATION OF A W-BAND REFLECTARRAY 37

of the LC inside each cell. Basically, the strategy relies on the switching-off speed of LCbeing longer than the refreshing period[32].

• Active addressing: It involves introducing an appropriate element with memory (capacitor,transistor) in each cell, so that it holds the charge for a limited period of time. Theswitching action of the external element helps the reflectarray cells to remain active untilthe next refreshing period [32].

5.2 Reflectarray Sample Under Test

The reflectarray antenna that needs to be characterized exploits the dielectric anisotropy of LCto produce a reconfigurable antenna. It has been designed to operate at 96 - 104GHz and it isexpected to generate about 440◦ phase variation and to introduce mean losses below 7 dB.

Figure 5.2 depicts the topology of the reflectarray. The unit cells are composed of threedipoles of different lengths that are printed on a quartz wafer, which is separated from ametal ground by an LC layer. 52 × 54 cells are arranged in a rectangular lattice to form thereflectarray [34].

(a) (b)

Figure 5.2: Schematic of the manufactured reflectarray [34]: (a) Unit-celland (b) reflectarray.

Table 5.1 summarizes the dimensions and the simulated results of the designedreflectarray [35].

Parameter Value

Operating frequency 96 - 104GHz

Operating incidence angle 0 - 30◦

Maximum reflection loss 12 dB

Phase variation at 100GHz 440◦

Parameter Value

Number of column cells 54 cells

Number of row cells 52 cells

Diameter of the reflectarray 60mm

Total diameter of the sample 100mm

Table 5.1: Specifications of the reflectarray sample.

38 5.3. QUASI-OPTICAL TEST BENCHES

Figure 5.3 presents a photograph of the manufactured reflectarray sample. As observed,the reflectarray cells are connected in rows and columns following a passive matrix addressingstrategy. A cell is biased if a potential difference is present between the row and column to whichit is connected. For independent excitation of each row and column, 52 + 54 biasing cables areextracted from the reflectarray and connected to the excitation source.

(a) (b)

Figure 5.3: Photographs of the reflectarray sample: (a) Front view and(b) rear view.

5.3 Quasi-optical Test Benches

This section presents the measurement set-ups that have been utilized for measuring thereflectarray for various angles of incidence. Quasi-optical set-ups for 30◦ and 45◦ incidenceangles have been proposed and analyzed using the software developed in Chapter 4:

• Quasi-optical set-up for 30◦ incidence: (Section 5.3.2).

– Based on lenses.

• Quasi-optical set-up for 45◦ incidence: (Section 5.3.3).

– Based on lenses.

– Based on mirrors.

All the utilized measurement systems are afocal systems (see Section 4.1.2) to which theVNA is connected. Therefore, the beam can be considered locally at the sample position as aplane wave regardless of the frequency.


It is also important to comment, that the calculated efficiencies are not representative of thereal behaviour of the system, since they are only considering spillover losses and losses in thelenses. Other sources of losses, dispersion or reflections could cause a loss in performance.

5.3.1 Utilized optical components

Only those optical components that are already available at the laboratory are utilized to designthe optical test benches:

1. Conical horns:

– 23 dB gain dual mode conical horns (×2).

2. Paraboloid off-axis mirrors:

– 45◦, ∅ = 101.6mm and Feq = 119.03mm (×2).

3. Dielectric lenses:

– Teflon, ∅ = 80mm and Feq = 62.7mm (×2) .

4. Supporting structures.

5. 600× 600 mm optical bench.

Available datasheets of these components can be found in Appendix G. Besides, the phasecenter of the horns has been calculated in Appendix C simulating them using HFSS (HighFrequency Structural Simulator R©), so that they can be placed at their optimum positionsaccording to the test bench designs.

5.3.2 Optical set-up for 30◦ incidence

Using the available material, the simplest test bench for measuring with 30◦ incidence consistsof placing the sample obliquely between two horns and collimating the beam with two lenses atthe output of the horns. Figure 5.4 illustrates the simulated behaviour of the beam through theproposed test bench at 94GHz, which is the frequency of interest for which the beam radius atthe sample is larger. Table 5.3 presents a summary of the obtained results for the proposed testbench.

• Edge diffraction is negligible since the amplitude taper at the sample and the rest of opticalstructures is below �17.4 dB.

• Spillover losses and other beam truncation effects can also be neglected since the edgetaper at every structure is sufficiently low. The main source of losses are the lenses whichintroduce 19.6% losses each.

• Lenses overlap, so in order to obtain a focused beam at the sample, their edges have beenslightly trimmed so that they fit in this configuration. This adjustment is not expected toperturb the behaviour of the lenses.


-50 0 50 100 150

(mm)

-100

-50

0

50

(mm

)Optical Bench Configuration

Beam e-1

Beam e-2

Figure 5.4: Test bench configuration for 30◦

incidence at 94GHz.

Optical element x(mm) y(mm)

Horn 1 (phase center) 0 0

Lens 1 (convex face) 69.1 0

Sample (center) 131.8 0

Lens 2 (convex face) 89.8 −72.8

Horn 2 (phase center) 65.9 −114.1

Table 5.2: Element positions for the proposedoptical test bench for 30◦ incidence.

Parameter 94GHz 100GHz

Beam at the sample

Beam radius 14.18mm 13.65mm

Radius of curvature ∞ m ∞ m

Efficiency

Edge taper at the sample�28.99 dB �31.28 dB

Maximum edge taper�19.94 dB �20.92 dB

Losses in the lenses 35.3% 35.3%

Spillover efficiency 98.35% 99.78%

Total efficiency ≈ 63.6% ≈ 64.5%

Table 5.3: Simulated results of the proposed test bench configuration tomeasure the reflectarray with an angle of incidence of 30◦.

5.3.3 Optical set-up for 45◦ incidence

According to the study developed in Section 4.3, there are different optical configurations thatcould produce satisfactory results for the measurement of the reflectarray sample with an angleof incidence of 45◦.

The simplest test bench again consists of utilizing two lenses to collimate the beam emergingfrom the horns. However, each lens introduces about 1 dB loss, so an alternative set-up thatuses off-axis mirrors as collimating elements has also been implemented. Empirical results ofboth measurement set-ups are compared in Section 5.3.4.


Proposed optical test bench using lenses

This optical set-up uses lenses to collimate the beam emerging from the horns. Figure 5.5illustrates the simulated behaviour of the beam through the proposed test bench at 94GHz,which is the the frequency of interest that produces the largest beam radius at the sample.Table 5.5 presents a summary of the obtained results for the proposed optical system.

-50 0 50 100 150 200

(mm)

-200

-150

-100

-50

0

50

(mm

)

Optical Bench Configuration

Beam e-1

Beam e-2


incidence at 94GHz using lenses.



Lens 1 (convex face) 47.7 0

Sample (center) 131.8 −69.1

Lens 2 (convex face) 131.8 −84.1


Table 5.4: Element positions for the proposedoptical test bench for 45◦ incidence using lenses.

• Edge diffraction is negligible since the amplitude taper at the sample and the rest of opticalstructures is below �17.4 dB.

• Spillover losses and other beam truncation effects can also be neglected since the edgetaper at every structure is sufficiently low. Therefore, the estimated loss in performanceis mainly due to the losses introduced by the lenses.

• The effective beam radius at the sample is larger than in the case of 30◦ incidence, becausethe sample is impinged more obliquely. Therefore, spillover losses are larger.



Beam at the sample



Spillover efficiency



Losses in the lenses 35.3% 35.3%


Total efficiency ≈ 63.8% ≈ 64.1%

Table 5.5: Simulated results of the test bench configuration to measurethe reflectarray with an angle of incidence of 45◦ using lenses.

Proposed optical test bench using 45◦ off-axis mirrors

In contrast to the previous optical configuration, this set-up is based on the use 45◦ off-axismirrors to collimate the beam emerging from the horns. Figure 5.6 illustrates the behaviour ofthe beam through the proposed test bench for a frequency of 94GHz, which is the frequencyof interest that produces the largest beam radius at the sample position. Table 5.7 summarizesthe obtained simulation results.

-50 0 50 100 150

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2


incidence at 94GHz using mirrors.



Lens 1 (convex face) 119 0

Sample (center) 34.9 −84.2

Lens 2 (convex face) 119 −168.3

Horn 2 (phase center) 0 −168.3

Table 5.6: Element positions for the proposedoptical test bench for 45◦ incidence using mirrors.

• Edge diffraction would be high at the borders of the sample, since they will be illuminatedwith about �5.5 dB with respect to the center of the sample.

• Spillover at the sample position will be large and it is the main reason for the loss inperformance observed in the estimated efficiency. Other truncation effects, such as beam


distortion, cannot be neglected, neither can they be evaluated using the implementedsoftware.

• Figure 5.6 suggests that spillover may travel towards the horns corrupting themeasurements. Indeed, the problem will be that the quartz frame in which the sampleis inserted is 40mm larger than the actual reflectarray. Therefore, the quartz will beilluminated and its response will partially obscure the reflectarray response.


Beam at the sample



Efficiency




Table 5.7: Simulated results of the alternative test bench configurationto measure the reflectarray with an angle of incidence of 45◦ using mirrors.

5.3.4 Comparison of lens-based and mirror-based set-ups for 45◦ incidence

As stated in Section 5.3.3, two different optical set-ups could yield good results for themeasurement of the reflectarray when it is impinged at an angle of 45◦. The first set-up isbased on lenses and the second is based on 45◦ off-axis mirrors.

Both proposed measurement systems have been implemented in the optical bench. In orderto decide which of them yields the best results, an optical thru (flat metallic plate) with thesame dimensions as the reflectarray is placed at the sample position. Photographs of the opticaltest benches utilized for these situations are presented in Figure 5.7.

(a) (b)

Figure 5.7: Optical thru in the quasi-optical test bench utilized tomeasure vertical incidence response for 45◦ incidence: (a) Based on lensesand (b) based on mirrors.


Figure 5.8 presents the response of the thru for both measurement systems in time and infrequency domains, where a rectangular window is used for the transformation and a Hammingfilter for time gating.

0 2000 4000 6000 8000 10000 12000

Time (ps)

0

10

20

30

40

50

60

(dB

)

Transmission coefficient in time domain

S21-LensesS21-Mirrors

(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-20

-15

-10

-5

0

(dB

)

Transmission coefficient in frequency domain after gating

Raw S21-LensesGated S21-LensesRaw S21-MirrorsGated S21-Mirrors

(b)

Figure 5.8: Measured S21 of the optical thru for 45◦ incidence: (a) Timedomain and (b) frequency domain.

Even though less reflections and lower losses were expected from the mirror-based test bench,a S21 response with more ripples and lower mean level has been actually obtained. This can beexplained understanding the multipath time response of Figure 5.8a:

• The earliest arriving signal corresponds to the response of the thru, since waves travelthrough the shortest path between both horns. This peak occurs latter for the mirror-basedsystem due to the larger effective focal distance of the mirrors.

• The rest of the peaks represent standing waves inside the system finally arriving to thehorn after several reflections. It can be seen that they arrive every certain period of timeaccording to the separation between optical elements in the test bench.

• Transmission losses are about 4 dB higher in the case of the mirror-based system, whichcould be explained from the high spillover losses that are occurring when the Gaussianbeam impinges at the sample (see Figure 5.6). However, spillover losses were includedin the calculation of the expected efficiency, which yielded worst performance for thelens-based system (see from Section 5.3.3).

• Another explanation is that in the case of mirrors, that energy is not lost; instead it arrivesin the form of those powerful “late-arriving signals”, whose power is more blurred for thelens-based system. This means that part of the loss in performance of the mirror-basedsystem is due to high return losses, which is an effect that was not not accounted for inthe efficiency estimation from Section 5.3.3.

As it is commented in Appendix D and Appendix E, the dynamic margin obtained aftercalibrating the VNA is about 25 dB, a quantity from which the losses occurring inside theoptical bench must be subtracted (losses of the thru). Therefore, when the mirror-based systemis used, high resonances of more than 20 dB appearing in the reflectarray response are actuallyfalling below the calibration precision, which corrupts the phase responses with noise and theycannot be recovered using the post-processing technique explained in Appendix E. For all thesereasons, the measurements of the reflectarray that are based on mirrors have been ignored forthe following analysis.


5.4 Reflectarray Characterization

This section presents the first set of measurements of the reflectarray sample described inSection 5.2. They are intended to obtain a general overview of the reflectarray response andvalidate the theoretical model and the manufacturing method.

5.4.1 Measurement plan

Reflectarrays behave very differently depending on how the cells are excited and thecharacteristics of the incident field. Therefore, the reflectarray behaviour can be characterizedvarying different parameters:

• For different incident frequencies.

• For different angles of incidence.

• For different polarizations.

• For different addressing strategies.

• For different signal shapes.

• For different control voltages.

• For different synthesized patterns.

• · · ·

The general parameters that are common for all the performed measurements are summarizedin Table 5.8.

Parameter Value

Range of frequencies 80 - 110GHz

Range of control voltages 0 - 300Vpp

Frequency of the excitation 1 kHz

Table 5.8: Common parameters to all the measurements.

Ranges of frequency and incidence angles for which to perform the measurements are chosenaccording to the specifications in Table 5.1, so that performance losses when being outside theoperational ranges can also be appreciated. The following list summarizes the measurementsthat are presented along the rest of the section:

• Quasi-static homogeneous control: Cells are all excited with the same quasi-staticcontrol signal and they are characterized for different biasing voltages (Section 5.4.2).

– 30◦ incidence for vertical orientation.

– 30◦ incidence for horizontal orientation.


• Dynamic control based on time-multiplexing: Short excitation pulses aresequentially applied to each row and column intending to maintain a certainquasi-permanent state in all the cells (Section 5.4.3):


– 30◦ incidence for horizontal orientation.

46 5.4. REFLECTARRAY CHARACTERIZATION

Section 5.3 presents the designs of the quasi-optical test benches that have been utilized tomeasure the reflectarray for both 30◦ (see Figure 5.4) and 45◦ incidence (see Figure 5.5). On theother hand, the VNA configuration and other measurement equipment is detailed in Appendix Dand applied post-processing techniques are explained in Appendix E.

5.4.2 Static homogeneous control

For this first set of measurements, every unit cell of the reflectarray is identically biased. Inorder to do this, the same quasi-static excitation has been simultaneously applied to every rowand column biasing lines. Measurements have been performed for combinations of:

– Vertical and horizontal orientation.

– 30◦ and 45◦ incidence angles.

– Sinusoidal and square excitation signals.

30◦ incidence for vertical orientation

A photograph of the reflectarray placed in the quasi-optical test bench is presented in Figure 5.9.

Figure 5.9: Reflectarray in the quasi-optical test bench utilized tomeasure vertical orientation response for 30◦ incidence.

The results obtained after applying the same sinusoidal excitation to every cells are presentedin Figure 5.10. Figure 5.11 shows the same experiment but for square excitation.

80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)

Sinusoidal excitation: |S21|

0.0 Vpp3.9 Vpp7.7 Vpp11.6 Vpp15.4 Vpp19.4 Vpp23.3 Vpp27.1 Vpp31.0 Vpp38.7 Vpp42.5 Vpp46.4 Vpp50.3 Vpp54.2 Vpp58.1 Vpp


(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Sinusoidal excitation: (S21)



(b)

Figure 5.10: Reflectarray measurements using sinusoidal excitation, 30◦

incidence and vertical orientation: (a) Amplitudes and (b) phase shifts.


80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)

Square excitation: |S21|

0 Vpp5 Vpp7.95 Vpp11.9 Vpp15.7 Vpp19.7 Vpp23.6 Vpp27.5 Vpp31.4 Vpp39.2 Vpp43.1 Vpp47 Vpp51 Vpp54.9 Vpp58.8 Vpp

63.6 Vpp67.4 Vpp71.3 Vpp78.4 Vpp87.3 Vpp95.2 Vpp103 Vpp112 Vpp119 Vpp129 Vpp137 Vpp145 Vpp153 Vpp223 Vpp307 Vpp

(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Square excitation: (S21)

0 Vpp5 Vpp7.95 Vpp11.9 Vpp15.7 Vpp19.7 Vpp23.6 Vpp27.5 Vpp31.4 Vpp39.2 Vpp43.1 Vpp47 Vpp51 Vpp54.9 Vpp58.8 Vpp

63.6 Vpp67.4 Vpp71.3 Vpp78.4 Vpp87.3 Vpp95.2 Vpp103 Vpp112 Vpp119 Vpp129 Vpp137 Vpp145 Vpp153 Vpp223 Vpp307 Vpp

(b)

Figure 5.11: Reflectarray measurements using square excitation for 30◦


30◦ incidence for horizontal orientation

In order to measure the response when the reflectarray is horizontally oriented, both the hornsand the sample are rotated by 90◦, so that TM polarization is maintained. A photograph of thereflectarray placed in the quasi-optical test bench is presented in Figure 5.12.

Figure 5.12: Reflectarray in the quasi-optical test bench utilized tomeasure vertical orientation response for 30◦ incidence.

Figure 5.13 presents the results that have been obtained applying the same sinusoidalexcitation voltage to every cell of the reflectarray. Figure 5.14 shows the results for squareexcitation.


80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)




(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1000

-800

-600

-400

-200

0

200

S21

(º)




(b)

Figure 5.13: Reflectarray measurements using sinusoidal excitation, 30◦

incidence and horizontal orientation: (a) Amplitudes and (b) phase shifts.

80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)




(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1000

-800

-600

-400

-200

0

200 S

21 (º

)Square excitation: (S21)



(b)


incidence and horizontal orientation: (a) Amplitudes and (b) phase shifts.


Photographs of the reflectarray placed in the proposed quasi-optical test benches are presentedin Figure 5.15. As concluded in Section 5.3.4, the set-up that yields the best results is the onebased on lenses. Therefore, the measurements taken using the optical set-up of Figure 5.15a arethe ones that are post-processed and presented in this section.

Figure 5.16 shows the results that have been obtained simultaneously applying the samesinusoidal excitation voltage to every cell of the reflectarray. Figure 5.17 presents the sameresults but for square excitation signals.


(a) (b)

Figure 5.15: Reflectarray in the quasi-optical test bench utilized tomeasure vertical orientation response for 45◦ incidence: (a) Based on lensesand (b) based on mirrors.

80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)



63.0 Vpp66.8 Vpp70.6 Vpp77.5 Vpp85.0 Vpp93.9 Vpp102.0 Vpp117.0 Vpp127.0 Vpp135.0 Vpp142.0 Vpp151.0 Vpp219.0 Vpp298.0 Vpp

(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0 S

21 (º

)



63.0 Vpp66.8 Vpp70.6 Vpp77.5 Vpp85.0 Vpp93.9 Vpp102.0 Vpp117.0 Vpp127.0 Vpp135.0 Vpp142.0 Vpp151.0 Vpp219.0 Vpp298.0 Vpp

(b)

Figure 5.16: Reflectarray measurements using sinusoidal excitation for45◦ incidence and vertical orientation: (a) Amplitudes and (b) phase shifts.

80 85 90 95 100 105 110

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)




(a)

80 85 90 95 100 105 110

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Square excitation: (S21)



(b)



Discussion of the results

This reflectarray was designed to work at 96 - 104GHz with an angular range of 0 - 30◦ andTM polarization. However, there are disagreements with respect to simulations that cannot


be explained from manufacturing tolerances, especially regarding the frequency band and themaximum phase difference (see Table 5.1). Therefore, the entire design and manufacturingprocess is being reconsidered [34].

0 50 100 150 200 250 300Voltage(Vpp)

-400

-300

-200

-100

0

Phas

e ra

nge

(º)

Sinusoidal excitation=30º, 94 GHz, H-Pol=30º, 100 GHz, H-Pol=30º, 94 GHz, V-Pol=30º, 100 GHz, V-Pol=45º, 94 GHz, V-Pol=45º, 100 GHz, V-Pol

(a)

0 50 100 150 200 250 300Voltage(Vpp)

-400

-300

-200

-100

0

Phas

e ra

nge

(º)

Square excitation=30º, 94 GHz, H-Pol=30º, 100 GHz, H-Pol=30º, 94 GHz, V-Pol=30º, 100 GHz, V-Pol=45º, 94 GHz, V-Pol=45º, 100 GHz, V-Pol

(b)

Figure 5.18: Comparison of the measured phase ranges at 94GHz and100GHz: (a) Sinusoidal excitation and (b) square excitation.

0 50 100 150 200 250 300Voltage(Vpp)

-25

-20

-15

-10

-5

0

S21(

dB)

Sinusoidal excitation

=30º, 94 GHz, H-Pol=30º, 100 GHz, H-Pol=30º, 94 GHz, V-Pol=30º, 100 GHz, V-Pol=45º, 94 GHz, V-Pol=45º, 100 GHz, V-Pol

(a)

0 50 100 150 200 250 300Voltage(Vpp)

-25

-20

-15

-10

-5

0

S21(

dB)

Square excitation

=30º, 94 GHz, H-Pol=30º, 100 GHz, H-Pol=30º, 94 GHz, V-Pol=30º, 100 GHz, V-Pol=45º, 94 GHz, V-Pol=45º, 100 GHz, V-Pol

(b)

Figure 5.19: Comparison of the measured losses at 94GHz and 100GHz:(a) Sinusoidal excitation and (b) square excitation.

A shift has occurred on the center frequency of the antenna, opening the possibility of usingthe reflectarray at 94GHz instead of at 100GHz. This can be observed in Figure 5.18 andFigure 5.19, which present, respectively, the obtained phase difference and amplitude loss fordifferent biasing voltages at 94GHz and at 100GHz.

• Phase linearity is approximately maintained in the range of 90 - 99GHz, although the phaserange was expected to be larger (about 440◦). At 100 - 104GHz, phase curves fall steeply,so small bias variations will cause large phase shift changes. This forbids the use of theantenna at this frequency band.

• Figure 5.18 shows that there is a threshold voltage from which the obtained phase shiftsaturates. Biasing voltages above the threshold will produce a faster response.

• Amplitude peaks observed in the measured S21 are caused by the different resonancefrequencies of the three dipoles of the cell. These peaks move when the excitation is


changed since resonance frequency of the cell varies, but they are excessively pronounced(< −20 dB) in the 100 - 104GHz band (see Figures 5.10a, 5.11a, 5.13a and 5.14a).

• Even though at 94GHz the measured range of phase shifts is shorter, the maximumamplitude losses are about 7 dB, in contrast to the more than 20 dB losses occurringfor certain permittivity values at 100GHz.

Maximum phase variation was expected to be about 440◦, but this value has significantlydecreased to 350 - 380◦, with horizontal orientation measurements showing slightly larger phasedifferences. Besides, as expected, when the angle of incidence is 45◦, the phase range is shorter,since 45◦ lies outside the operating angular range (0 - 30◦).

It can also be mentioned that a square excitation produces a phase shift range and amplituderesponse similar to that of a sinusoidal excitation, but displaced in the voltage dimension. Thisis compliant with the developed LC model, which shows that the effective permittivity is amonotonically increasing function of the RMS (Root Mean Square) voltage [34].

Considering all of the above, it has been concluded that this reflectarray prototype canbe used for pattern synthesis experiments in the 92 - 96GHz band and for both vertical andhorizontal 30◦-incidence. Array pattern synthesis requires that each element of the reflectarraycan be independently biased to produce all possible phase shifts from 0◦ to 360◦. At 94GHz, themeasured phase-shift variation barely reaches 240 - 270◦, so the 360◦ can not be covered. Thesolution to cover the entire 360◦-range without actually generating the highest phase-shift valuesconsists of quantizing the states so that only a finite number of phase shifts are generated. Inthis case, a 270◦ phase variation allows to generate four maximally-separated states (0◦ ± 45◦,90◦ ± 45◦, 180◦ ± 45◦ and 270◦ ± 45◦), which is equivalent to consider each reflectarray cell asan electronically-controlled two-bit phase-shifter. However, using a small number of finite phaseshifts comes at the expense of higher side-lobe levels in the synthesized patterns [30]

Generating those four phase-shift states will be the goal of Section 5.4.3. In thatsection, instead of polarizing the reflectarray cells with different biasing voltages, the effectivepermittivity is changed by rapid discontinuous excitations that control the dynamic transientresponse of the liquid crystal cells and maintain them in intermediate excitation states.

5.4.3 Dynamic control based on time-multiplexing

The reflectarray sample under test utilizes a passive addressing strategy to excite each cell(see Section 5.1.2). This technique is based on the fact that the decay times (on-to-off time)of the LC are slower than its rising times (off-to-on time). Therefore, a strategy based ontime-multiplexed excitations over rows and columns can be applied to maintain each cell of thereflectarray independently biased on a certain state.

It should be noticed, that cells on one row or column will only be receiving the necessaryexcitation to maintain the maximum voltage a fraction of the time. Therefore, LC will actuallypresent an oscillating state in between the ON response (maximum voltage) and the OFFresponse (0Vpp) [32]. Knowing the dynamic behaviour of the cells (transient response of the LC),the pulse duration and the refreshing period can be tuned to generate different quasi-permanentstates [33].

Two different strategies that apply pulses of a certain fixed voltage and duration over thedifferent columns (one-dimensional sweep) are tested along this section. Their most importantparameters are summarized in Table 5.9.


Strategy 1: This excitation strategy has been developed by analyzing the dynamic behaviourof the LC cells according to the theoretical model [33]. Pulses of 300Vpp are appliedsequentially to the 52 columns. Maximum phase shift state uses the minimum possiblerefreshing period, which is 40ms/column × 52 columns = 2.08 s. As it will be observedin the measurements, this strategy produces very unstable states due to the high peakvoltage.

Strategy 2: This second excitation strategy is based on an empirical adjustment of theexcitation parameters. Besides, in order to reduce the oscillations, the columns at theedges of the reflectarray are grouped and excited together, so that the equivalent numberof columns in which to iterate is only 36. This allows to reduce the minimum refreshingperiod to 1.44 s. Furthermore, the excitation voltage is 225Vpp so that rising and decayingtimes are slower and fluctuations decrease.

VoltageExcitation duration / Refreshing period / Initial decay time

State 1 State 2 State 3 State 4

#1 300Vpp 0ms/–/– 17ms/8.32 s/20.8 s 13ms/8.32 s/6.24 s 40ms/2.08 s/0 s

#2 225Vpp 0ms/–/– 17ms/5.76 s/14.4 s 25ms/5.76 s/4.32 s 40ms/1.44 s/0 s

Table 5.9: Summary of both excitation strategies for dynamictime-multiplexed control of the reflectarray states.

The measurements presented within this section attempt to characterize the dynamicbehaviour of the reflectarray cells when a time-multiplexed signal is used to maintain themall in the same orientation state. Since the responses oscillate, several captures have been takenfor each state to observe their stability once the permanent regime of each particular state hasbeen reached.

The ultimate goal of these measurements is to find four excitation patterns that producefour different quasi-permanent reflectarray states (as a 2-bit phase-shifter) at 94GHz and30◦-incidence, for both antenna orientations.


The measurement set-up for this configuration was presented in Figure 5.9. The excitationstrategy excites every cell periodically according to the theoretical model (strategy 1).Measurement results are presented in Figure 5.20.


88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)

Dynamic states: amplitude

0 VppState 2State 3State 4300 Vpp

(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Dynamic states: phase shift


(b)

Figure 5.20: Reflectarray performance for the four analyzed dynamicstates for vertical orientation (strategy 1): (a) Amplitudes and (b) phaseshifts.

30◦ incidence for horizontal orientation

The measurement set-up for this configuration was presented in Figure 5.12. In this case, bothstrategies have been tested:

• The results for the first strategy are presented in Figure 5.21.

• The results for the second strategy are presented in Figure 5.22.

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)



(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)



(b)

Figure 5.21: Reflectarray performance for the four analyzed dynamicstates for horizontal orientation (strategy 1): (a) Amplitudes and (b) phaseshifts.


88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-40

-35

-30

-25

-20

-15

-10

-5

0

S21(

dB)



(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)



(b)

Figure 5.22: Reflectarray performance for the four analyzed dynamicstates for horizontal orientation (strategy 2): (a) Amplitudes and (b) phaseshifts.

5.4.4 Discussion of the results

Different states of the reflectarray have been obtained using time-multiplexed excitations andtheir mean phase shifts can be observed in Figure 5.23.

• The first strategy consisted of exciting each row and column with the same period andduring the same time. However, the mean phase shift values that have been obtained arenot sufficiently separated to apply them for 2-bit-based pattern synthesis.

• A second approach utilized an empirical adjustment of the excitation parameters, sothat the mean obtained phase shifts were maximally separated and more stable (seeFigure 5.23c).

The frequency for which the excitation strategies were designed is 94GHz and fourwell-differentiated states were expected.

• For vertical and horizontal orientation, the strategy based on the theoretical modelproduces two states that are almost equal in average, so indeed there are only 3 states.Errors in the theoretical model, the manufacturing process or the excitation circuitry arebeing considered.

• The second strategy has only been applied for horizontal orientation, but it achieves fourwell differentiated states that are separated by about 80◦. These results indicate that it ispossible to use this passive addressing approach to obtain different quasi-permanent phaseshifts. If different states could be independently applied to different cells, the reflectarraycould be used to synthesize various radiation patterns.

• In the case of horizontal orientation, Figure 5.23 shows that phase responses are moredifferent at 96GHz than at 94GHz. This could already be seen in the static analysis fromFigure 5.14.

Oscillations on the phase and amplitude responses are too large in comparison to what itwould be required (±45◦).


88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Dynamic states: mean phases (V-pol)

0 VppState 2State 3State 4

(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Dynamic states: mean phases (H-pol)


(b)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-1200

-1000

-800

-600

-400

-200

0

S21

(º)

Dynamic states: mean phases (H-pol)


(c)

Figure 5.23: Mean values of the phase curves for the different dynamicexcitation strategies: (a) vertical orientation (strategy 1), (b) horizontalorientation (strategy 1) and (c) horizontal orientation (strategy 2).

• Using the first excitation strategy, vertical orientation produces states that oscillate morethan 75◦. Horizontal orientation states oscillate more than 130◦.

• The second addressing strategy, as expected, produces less oscillations due to the reductionin the refreshing period and the peak voltage of the excitation pulses.

Finally, since a limited number of captures were randomly taken at different excitationinstants, some oscillation effects may not have been observed. Results will be more representativewhen captures are taken automatically several times within a refreshing period, which is plannedto be done in the future.

Table 5.10 presents a summary of the results obtained for the measurements performedwithin this section, at 94GHz for vertical orientation and at 96GHz for horizontal orientation.


Parameter State 1 State 2 State 3 State 4

Vertical orientation (strategy 1)

Mean relative phase shift 0◦ �118◦ �98◦ �219◦

Phase shift variation — 76◦ 71◦ 45◦

Mean amplitude �5.7 dB �9.7 dB �9.9 dB �4.7 dB

Amplitude variation — 1.6 dB 0.7 dB 2 dB

Horizontal orientation (strategy 1)



Mean amplitude �4.8 dB �9.4 dB �9.5 dB �6.9 dB

Amplitude variation — 2.3 dB 2.3 dB 3.6 dB

Horizontal orientation (strategy 2)



Mean amplitude �4.6 dB �8.3 dB �9 dB �6.9 dB

Amplitude variation — 1.2 dB 3.6 dB 3 dB

Table 5.10: Summary of the reflectarray performance for the analyzeddynamic states at 94GHz for vertical orientation and at 96GHz forhorizontal orientation.

5.4.5 Future measurements

There are still several features of the reflectarray performance that need to be characterized.These tasks are intended to be performed in the future and are not included in this document.

The first important measurement that needs to be performed is the characterization of thedynamic behaviour of the reflectarray cell. Since the measurements of Section 5.4.3 have notproduced the expected theoretical results, a more accurate model of the the rising (off-to-ontime) and decaying (on-to-off time) speed of the LC is needed. The easiest way to obtain thisis to configure the VNA to take periodic captures of the S-parameters during the transitionalperiods after switching on and off the excitation.

Once four different phase shifts are obtained using the time-multiplexed technique, differentradiation patterns could be synthesized. This would consist of applying a different biasing stateto each cell of the reflectarray. LC molecules of a reflectarray cell would maintain differentorientation states if their excitation is refreshed according to their required state. Crosstalkbetween nearby cells may occur, complicating the synthesis.

Finally, once different excitation strategies have been validated, the reflectarray will bemeasured again in an anechoic chamber, in which the environment is more controlled and moreaccurate results could be obtained.

Chapter 6

Radar Receiving ChainCharacterization.

This chapter addresses the characterization of the millimeter-wave receiving subsystem of theradar. The ultimate goal of is to determine whether it is possible to build the receiving chainof the space debris radar using the waveguide components that are already available at theMicrowave and Radar Group and to analyze the limitations and requirements of this choice.

In particular, the following measurements have been performed:

• S-parameters characterization of active and passive devices (Section 6.2).

• S-parameters characterization of the isolation chain of the receiver (Section 6.3).

• Noise characterization (Section 6.4).

The experimental acquisitions obtained during the measurements will be post-processedusing Matlab and ADS (Advanced Design System R©).

6.1 The Millimeter-Wave Receiving Subsystem

Considering the high cost of waveguide components, one of the main criteria for the design ofthe architecture was the utilization of W-band components that are already available at thelaboratory [3]. The complete radar architecture was presented in Figure 2.1 and that basicdesign was modified in Figure 2.2 to include monopulse tracking capabilities.

The main goal of the receiving subsystem of the radar is the amplification of the receivedsignal and its downconversion to a band from which it can be digitalized. On the other hand, inorder to maintain a low noise figure, an LNA is introduced at the input of the receiving chain, sothat it establishes the noise floor of the receiver [7]. However, since this a monostatic radar thatuses a single antenna for transmitting and receiving, the receiver must be switched-off duringthe time the transmitter is active to protect the devices of the receiver from being damaged bythe high transmitted power. For this purpose, an isolation chain is introduced before the LNA[7]. This will degrade the overall noise figure, since the LNA no longer is the first component ofthe receiver. There are two different proposals for this isolation chain:

Circulator-based: This option is presented in Figure 6.1a. It is formed by a circulator,

57

58 6.2. CHARACTERIZATION OF INDIVIDUAL COMPONENTS

an isolator and a switch. It is the simplest option but it can only be used with thenon-monopulse architecture.

OMT-based: This option is presented in Figure 6.1b. Isolation is procured by an OMT(Orthomode Transducer) followed by an isolator and a switch. This combination canbe used in both monopulse and non-monopulse architectures (see Figure 6.1c).

mm-wave receiving

subsystem

Isolator

AntennaHPA

LNA

Switch

MixerLO

Active IF filter

CirculatorCCiirrccuullaattoorr

(a)

mm-wave receiving

subsystem

Isolator

AntennaHPA

LNA

Switch

MixerLO

Active IF filter

OMT

(b)

mm-wave receiving

subsystem

Isolator

AntennaHPA

LNA

Switch

LO

Mixer

Active IF filter

2OMT

Isolator

LNA

Switch

Mixer

Active IF filter

Divider

(c)

Figure 6.1: Millimeter-wave receiving subsystem options: (a)Circulator-based and (b) OMT-based and (c) OMT-based for monopulse.

The following sections will characterize the proposal of Figure 6.1a, since the OMT ofFigure 6.1c is being designed together with the monopulse horn and is not yet available. Indeed,the monopulse horn has been specified to provide 6 dB higher isolation between ports than thecirculator, so it will yield better performance if the manufacturing process is correct.

6.2 Characterization of Individual Components

Appendix F presents a detailed characterization of some of the available millimeter-wavecomponents that may be used in the radar if their response is considered appropriate for theapplication. The specific devices that are proposed to be utilized in the receiver subsystem fromFigure 6.1a are listed below:

• Circulator: ELVA CR-1094 (Section F.3)

• Isolator: RPG WFI-110 (Section F.1)

• Switch: ELVA SPST-10 (Section F.5)

• LNA: RPG W-LNA75110 (Section F.7)

• Mixer: Quinstar QMB-9999WS balanced mixer.

• Active base band filter: dependent on ADC requirements.

CHAPTER 6. RADAR RECEIVING CHAIN CHARACTERIZATION. 59

Table 6.1 summarizes the performance of those components from the list whose S-parametershave been measured in Appendix F. The manufacturers’ datasheets corresponding to each devicecan be found in Appendix G.

Measured Datasheet

Circulator

Insertion losses 0.22 dB 0.7 dB

Return losses 26.7 dB 20.8 dB

Isolation 25 dB 23 dB

Bandwidth[1] 7GHz 2.5GHz

Isolator

Insertion losses 1.6 dB 1.3 dB

Return losses 19.7 dB 14 dB

Isolation 22.9 dB > 20 dB

Switch

Insertion losses 0.96 dB 1 dB


Return losses (OFF) 0.55 dB —

LNA

Gain 28 dB 26.5 dB

Return losses (input) 9.2 dB 9.3 dB

Table 6.1: Summary of the measured responses of the receiving chainmillimeter-wave components at 94GHz.

6.3 Characterization of the Receiver Isolation Chain

As commented in Section 6.1, monostatic pulsed systems require an isolation chain to protectthe receiver. This section will show the results obtained from the S-parameters characterizationof the isolation scheme proposed in Figure 6.1a.

The switch will be off-biased when the transmitter is active to protect the receiver and it willbe on-biased during the time in which the transmitter is inactive to minimize receive insertionlosses.

There are four different situations that need to be analyzed:

1. Power transfer from the transmitter to the antenna (transmit mode, OFF): Figure 6.2a.

2. Power transfer from the transmitter to the receiver (transmit mode, OFF): Figure 6.2b.

3. Power transfer from the antenna to the receiver (transmit mode, OFF): Figure 6.2c.

4. Power transfer from the antenna to the receiver (receive mode, ON): Figure 6.2c.

[1]The circulator response is actually what limits the bandwidth of the receiver.

60 6.3. CHARACTERIZATION OF THE RECEIVER ISOLATION CHAIN

(a) (b) (c)

Figure 6.2: Important situations that need to be analyzed: (a) Insertionloss in transmit mode, (b) transmit-receive isolation in transmit mode and(c) insertion loss in transmit and receive modes.

6.3.1 Transmit insertion losses.

Power transfer from the antenna to the transmitter is given by the insertion losses of thecirculator when it is loaded at its third port by the receiving chain. This situation can beobserved in Figure 6.2a.

Considering that input return losses of the isolator are larger than 18.8 dB for the band ofinterest, transmit insertion losses will be very similar to the insertion losses of the circulatoralone, which are presented in Figure F.10a.

A summary of the obtained results when power is flows from the transmitter to the antennais presented in Table 6.2.

Parameter Measured

Insertion losses (S21) 0.15 dB

Return loss (S11) 26.7 dB

Isolation (S12) 36 dB

Bandwidth 91 - 98GHz

Table 6.2: Transmitter-to-antenna performance at 94GHz.

6.3.2 Isolation between the transmitter and the receiver.

Power transfer from the transmitter to the receiver should be as low as possible in order toprevent damages on the receiver when the transmitter is active. The schematic for this situationis presented Figure 6.2b.

Figure 6.3 shows the measurement set-up that has been utilized for the characterization ofthis ensemble. Since the horn is not available, the port where the antenna should be connectedis matched with a load (characterized in Figure D.2).


Figure 6.3: Measurement set-up for the isolation chain collocating theload at port 2 of the circulator.

Measured S-parameters for each state of the switch are shown in Figure 6.4. Even thoughthe behaviour of both states of the switch has been characterized, the receiver will always beswitched-off during transmission for its own protection, so only the off-state is relevant.

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (TX-OFF)

S11S13S31S33

(a)

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (TX-ON)

S11S13S31S33

(b)

Figure 6.4: Isolation chain performance from the transmitter to thereceiver: (a) S-Parameters amplitude when the switch is OFF and (b)S-Parameters amplitude when the switch is ON.

Insertion and return losses for each state are presented in detail in Figure 6.5, and the mostrelevant results obtained for this configuration are summarized in Table 6.3.

• Figure 6.5a shows that insertion losses when the switch is off-biased are larger than 60 dB[2]

in the band of interest (92 - 96GHz). This high isolation is produced by the circulator andthe off-biased switch.

• Obtained input return losses (Figure 6.5b) are larger than 22 dB in the band of interestand they are imposed by the return losses of the circulator (see Figure F.10b).

[2]Measurements below 40 dB are considered to be unreliable due to the finite precision of the calibration.

62 6.3. CHARACTERIZATION OF THE RECEIVER ISOLATION CHAIN

90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

20

30

40

50

60

70

IL (d

B)

Insertion Losses (TX to RX)

Total (OFF)Total (ON)

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

20

25

30

35

40

45

50

RL

(dB

)

Input Return Losses (TX to RX)


(b)

Figure 6.5: Isolation chain performance from the transmitter to thereceiver: (a) Insertion losses and (b) return losses.

Parameter Measured

Insertion losses (S31 OFF) > 60 dB

Return loss (S11 OFF) 23.7 dB

Table 6.3: Transmitter-to-receiver performance results at 94GHz.

6.3.3 Power transfer from the antenna to the receiver.

Power transfer from the antenna port to the LNA should be maximum during the periods oftime in which the receiver is listening to the echoes and minimum during the periods in whichthe transmitter is active. The schematic for this situation can be observed in Figure 6.2c.

Figure 6.6 shows the measurement set-up that has been utilized for the characterization ofthis chain of devices. The port that would be connected to the transmitter is matched with aload and S-parameters are measured from port 2 of the circulator to the output of the switch.

Figure 6.6: Measurement set-up for the isolation chain collocating theload at port 1 of the circulator.

• During the periods of time is which the transmitter is inactive, the switch must be on-biasedin order to minimize insertion losses.

• During the periods of time in which the transmitter is active, the switch must be off-biased


to increase isolation between the transmit and receive subsystems. The power originatedat the transmitter will not only reach the receiver directly through the circulator, but alsoafter reflecting at the antenna and going through the circulator again.

• During a small period of time after the transmitter is deactivated, the switch will remainoff-biased to prevent damages due to powerful early echoes (blanking).

Therefore, both switch positions are relevant and require characterization. MeasuredS-parameters for each state of the switch are presented in Figure 6.7.

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (RX-OFF)

S22S23S32S33

(a)

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (RX-ON)

S22S23S32S33

(b)

Figure 6.7: Isolation chain performance from the antenna port to theLNA port: (a) S-Parameters amplitude when the switch is OFF and (b)S-Parameters amplitude when the switch is ON.

Insertion and return losses for each state can be observed in detail in Figure 6.8, and themost relevant results have been summarized in Table 6.4.

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

5

10

15

20

25

30

35

40

IL (d

B)

Insertion Losses (Antenna to RX)


(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

15

20

25

30

35

40

45

RL

(dB

)

Input Return Losses (Antenna to RX)


(b)

Figure 6.8: Isolation chain performance from the antenna port to theLNA port: (a) Insertion losses and (b) return losses at the antenna port.

• Figure 6.8a shows that insertion losses of the isolation chain (receiving mode) areapproximately constant and below 2.8 dB in all the band of interest (92 - 96GHz). Thisis consistent with the expected 0.7 + 1.3 + 1 = 3dB insertion losses extracted from thedatasheets (see Appendix G).

64 6.4. NOISE PERFORMANCE OF THE RECEIVER CHAIN

• Total isolation (transmit mode) from the power coming from the antenna is larger than32 dB in the band of interest (Figure 6.8a). Therefore, transmitted power reflected atthe antenna or at early targets will be attenuated by that value. Indeed, the analysis ofSection 6.5 will show that reflections at the antenna are the main limitation to protectthe receiver. This means that, if the antenna is damaged or improperly connected, thereceiver might suffer irreparable damage.

• Obtained input return losses (Figure 6.8b) are larger than 22 dB in the band of interest.They are equal for both on and off switch states and they are imposed by the input returnlosses of the circulator (see Figure F.10b).

Parameter Measured

Insertion losses (S32 ON) 2.2 dB

Return loss (S22 ON) 23.7 dB

Isolation (S32 OFF) 35.6 dB


Table 6.4: Antenna-to-receiver performance results at 94GHz.

6.4 Noise Performance of the Receiver Chain

This section deals with the calculation (Section 6.4.1), simulation (Section 6.4.2) andmeasurement (Section 6.4.3) of the noise power level at the input of the digitalizer. In order todo this, the noise internally generated by the proposed receiving chain must be estimated. Theschematic for the complete receiver is presented in Figure 6.9.

Figure 6.9: Block diagram of the receiver.

The noise power spectral density of a system must be integrated across the receiverbandwidth. In the case of a radar system, the receiver bandwidth is equivalent to the bandwidth


of a bin of the FFT (Fast Fourier Transform), which should be approximately equal to theinverse of the pulse width (assuming rectangular windowing). Since the pulse duration is yetto be selected, the noise performance of several receiver bandwidths will be characterized alongthis section.

In order to guarantee that the receiver noise floor is not being limited by the digitalizer noisefloor, it has been necessary to include an active filter as the last component of the receiver (seeSection 6.4.1).

6.4.1 Analytical estimation using Friis formula

Friis formula is the easiest and most direct analytical approach to estimate the receiver noiseperformance. However, it does not take into account mismatches between components [36] andis not valid in this case due to the lack of an image rejection strategy. Therefore, an alternativeformulation has been derived [37].

The equivalent noise temperature at the output of the mixer referenced to the input of thechain is calculated in Equation 6.1:

Te = T circe +

T isole

Gcirc+

T swe

GcircGisol+

TLNAe

GcircGisolGsw+

Tmixe

GcircGisolGswGLNA(6.1)

Assuming that the noise of the desired and the image bands are added incoherently in the IFband, the noise at the output of the mixer can be calculated with Equation 6.2. Both sidebandswill be very close together (IF is very low), so it has been assumed that they have the same gainand the same noise figure.

N0mixer = kB(TSe + TS

a )GS + kB(T I

e + T Ia )G

I ≈ k2B(Ta + Te)G (6.2)

– TSe and T I

e are the equivalent noise temperatures of the mixer at the desired and imagebands.

– TSa and T I

a are the input noise temperatures at the desired and image bands.

– GS and GI are the gains at the desired and image bands from the first component of thechain up to the mixer.

Equation 6.3 defines an equivalent SSB (Single-Side Band) noise figure of the chain from thefirst component up to the mixer, that can be directly used in the classical Friis formulation [37].It is exactly 3 dB higher than the original DSB (Double-Side Band) noise figure [3].

FSSB |Ta=SNRin

SNRout=

Nout

NinG=

k2B(Ta + Te)G

kBTaG= 2 +

2Te

Ta= 2FDSB |Ta (6.3)

By using the adaptation from Equation 6.3, the noise figure of the chain and its equivalentnoise temperature can be calculated. The necessary parameters to calculate Friis formula can befound in Table 6.5. These have been obtained from measurements (Appendix F) and datasheets(Appendix G).

[3]In Equation 6.3 and the following derivations, noise figures are referenced to the antenna temperature Ta

instead of to the reference temperature T0 = 290K. This permits using Equation 6.7 regardless of the antennatemperature.


ComponentNoise figure for

T0 = 290KGain

Circulator 0.22 dB �0.22 dB

Isolator 1.6 dB �1.6 dB

Switch (ON) 0.96 dB �0.96 dB

LNA 3.8 dB 28 dB

DSB Mixer 6.5 dB �6.5 dB

Table 6.5: Friis formula parameters at 94GHz.

The noise figure of passive components is considered to be equal to their attenuation, whichis valid when the reference temperature T0 is equal to the room temperature Tamb. Besides,the DSB noise figure of the mixer has been assumed to be equal to its conversion losses, sinceits datasheet does not provide this information. These conversion losses are not entirely due tothermal noise, so the real value for the mixer noise figure is likely to be lower [38].

Equations 6.4, 6.5 and 6.6 present, respectively, the noise figure, equivalent temperature andtotal available gain of the receiver chain.

FSSB |T0= 9.59 dB (6.4)

Te = 2058.9K (6.5)

Gtotal = 18.72 dB (6.6)

Since the antenna is going to be pointed at the sky, antenna noise temperature will beassumed to be Ta = 60K. Besides, the output noise for the case in which the antenna noisetemperature is Ta = 290K has also been calculated, for the purpose of comparing this resultwith the measurements described in Section 6.4.3. Equation 6.7 can be used to calculate thenoise at the output of the mixer for both cases.

Nout = kTaF |Ta BRXGtotal (6.7)

The obtained values (Table 6.6) are below the digitalizer quantization noise, which (accordingto models under consideration: LTM9004 and AD9655) will introduce about �100 dBm in abandwidth of 40 kHz. In order to increase the dynamic range of the receiver, a 20 dB gainamplifier will be included at its output. This amplifier will not have much effect on the receivernoise figure because it is the last element of the chain. However, it will increase the overall gainof the system and therefore, the total output noise. Assuming that the amplifier design obtainsa noise figure better than 4 dB, the new noise figure will be FSSB |T0

= 9.61 dB, which is almostequivalent to the value obtained in Equation 6.4.

Table 6.6 presents the noise power levels obtained for different receiver bandwidths, withand without IF amplifier. It can be noticed that the obtained output noise power after theintroduction of the 20 dB active filter is more than 10 dB above the noise floor of the digitalizer.


Receiverbandwidth

Pulsewidth

Output noise (withoutIF amplifier)

Output noise (with IFamplifier)

Ta=60K Ta=290K Ta=60K Ta=290K

39 kHz 26 �s �100.15 dBm �99.75 dBm �80.13 dBm �79.74 dBm



Table 6.6: Output noise results at 94GHz for different receiverbandwidths obtained using Friis formula.

6.4.2 Noise budget analysis

Friis formulation does not take into account the effects of mismatches between the receivercomponents. A noise budget simulation has been performed in ADS, in order to obtain a betterestimation of the real noise figure and output noise power level of the receiver subsystem.

Figure 6.10 shows the schematic of the receiver chain that has been drawn in ADS in orderto carry out the simulation at 94GHz. RF frequency has been chosen to be 2MHz above theLO frequency in order to simulate a target producing a beat frequency of 2MHz. Besides, thesimulation has already been performed assuming the use of a 20 dB-gain IF amplifier at theoutput of the chain.

RF Input

Noise Figure Budget Analysis

Component and input/output noise figure

VarEqn

Ref

1 2

Ref

1 2

Ref

1 2

Amplifier2MixerWithLOAmplifier2S2PS2PS2P

Term

P_1Tone

VAR

AMP_IFMIXERLNASwitchIsolatorCirculator

Term2

PORT1

VAR1

Ta=290

LOfreq=94 GHz

S12=0S22=polar(0,180)S11=polar(0,0)S21=dbpolar(20,0)

ConvGain=dbpolar(-6.5,0)DesiredIF=RF minus LO

RBW=39 KHz

ZRef=50 OhmS21=dbpolar(28,0)

S12=0

S11=dbpolar(-9.2,0)S22=dbpolar(-9.2,0)

IFfreq=RFfreq-LOfreq

Temp=27Noise=noZ=50 OhmNum=2

Temp=Ta-273Noise=yesFreq=RFfreqP=dbmtow(Power_RF)Z=50 OhmNum=1

RFfreq=94.002 GHzPower_RF=-85

RF IF

Figure 6.10: Receiver schematic for the ADS noise budget analysis.

• Obtained noise figure is Ftotal = 9.59 dB, which is approximately equal to the resultobtained using the modified Friis equation from the previous section.

• As expected, Figure 6.11a shows how the ADS and modified noise figure calculation areabout 3 dB above the classical Friis formula result (see Figure 6.11a).

• The obtained output noise is about 2 dB larger than the one that would be obtained just byapplying the noise figure definition (Nout = kT0BFG). Instead, ADS noise calculation istaking into account mismatches between components in order to calculate a more realisticoutput noise value. Noise at the output of each receiver component can be observed inFigure 6.11b.


Circulator Isolator Switch LNA Mixer IF-AMP0

2

4

6

8

10

NF

(dB

)Simulated Noise Figure

ADS noise budgetFriis formulaModified Friis formula

(a)

Circulator Isolator Switch LNA Mixer IF-AMP-150

-140

-130

-120

-110

-100

-90

-80

-70

Nou

t(dB

m)

Simulated Output Noise Power

RBW=39KHz-Ta=290KRBW=24KHz-Ta=290KRBW=10KHz-Ta=290KRBW=39KHz-Ta=60KRBW=24KHz-Ta=60KRBW=10KHz-Ta=60K

(b)

Figure 6.11: Noise budget analysis of the receiver: (a) noise figure referredto 290K and (b) output noise power for Ta=290K.

Figure 6.11b and Table 6.7 summarize the results obtained for the different receiverbandwidths that have been considered along the simulations.

Receiverbandwidth

Pulsewidth

Output noise(after IF amplifier)

Ta=60K Ta=290K

39 kHz 25.6 �s �78.74 dBm �77.92 dBm

24 kHz 41.7 �s �80.85 dBm �80.03 dBm

10 kHz 10 �s �84.65 dBm �83.83 dBm

Table 6.7: Noise budget analysis results at 94GHz at the output of theIF active filter for different receiver bandwidths.

6.4.3 Noise measurements

Noise measurements of the receiving chain have been performed following the technique andconfigurations explained in Appendix D.

Figure 6.12 shows the particular set-up that has been used for measuring the noise at theoutput of the receiver chain. This set-up already includes a 20 dB-gain IF amplifier at its outputto maximize the dynamic range, although a new amplifier will be designed in the future. The IFfrequency in which the system will work is yet to be decided (0 - 1GHz), but there will be no bigdifferences on the results as long the mixer is working at its operating band. Oscillator frequencyis generated by multiplying by 6 a 16.667GHz signal produced with a signal generator.


Figure 6.12: Measurement set-up for the characterization of the receiverchain noise performance.

As previously commented, the receiver bandwidth is given by the width of an FFT bin,which is approximately equal to the inverse of the pulse duration. Since the RBW (ResolutionBandwidth) of the spectrum analyzer is the bandwidth over which power is integrated, thisis the parameter that is modified in order to characterize the performance for different pulsedurations. The obtained measurement results for each of the receiver bandwidths that havebeen characterized are presented in Figure 6.13.

• Figure 6.13 shows that noise is quite flat, without noticeable spurious signals, at least inthe range 0.5 - 5MHz.

• Measured noise power levels are about 1 dB higher than what was expected from the noisebudget simulation in ADS (see Table 6.7).

• The power decay occurring at 5MHz is due to the active filter (IF amplifier) at the outputof the mixer. The future IF amplifier will have at least 50MHz bandwidth.

1 2 3 4 5 6 7 8 9 10

Frequency (MHz)

-120

-110

-100

-90

-80

-70

P out(d

Bm

)

Noise traces at the filter output (LO=7dBm)

RBW=39KHz - LO OFFRBW=39KHz - LO ONRBW=24KHz - LO OFFRBW=24KHz - LO ONRBW=10KHz - LO OFFRBW=10KHz - LO ON

Figure 6.13: Noise power measurement at the output of the receiver chainwhen the transmitter and the antenna are matched (Ta = 290K).

Assuming that the loads are perfectly matched resistors (T = T0), the degradation on theoutput SNR (Signal to Noise Ratio) due to the introduction of the receiver components can beestimated from the measurement. In all the cases, the noise level curve (LO ON traces) is well


above the noise floor (LO OFF traces). This guarantees that the impact of the noise internallygenerated by the spectrum analyzer is negligible with respect to the measurement (excepting atvery low frequencies). A summary of the obtained results is shown in Table 6.8.

Receivebandwidth

Pulse widthOutputnoise at1MHz

Outputnoise at2MHz

Degradation

39 kHz 25.6 �s �76.8 dBm �76.8 dBm 12.57 dB

24 kHz 41.7 �s �78.8 dBm �78.9 dBm 12.58 dB

10 kHz 10 �s �82.7 dBm �82.7 dBm 12.58 dB

Table 6.8: Noise power level at the output of the receiver chain for anantenna temperature of Ta ≈ Tamb ≈ 290K.

6.4.4 Conversion losses

The datasheet of the multiplier (see Appendix G) indicates that the optimum input power valuefor this device is 7 dBm. Utilizing the same measurement set-up of Figure 6.12, the output powerof the signal generator has been swept in order to characterize its impact on the downconversionprocess.

1 2 3 4 5 6 7 8 9 10

Frequency (MHz)

-120

-110

-100

-90

-80

-70

P out(d

Bm

)

Noise traces at the filter output (RBW=10KHz)

LO=0dBmLO=2dBmLO=4dBmLO=6dBmLO=7dBmLO=8dBmNoise floor

(a)

1 2 3 4 5 6 7 8 9 10

Frequency (MHz)

-120

-110

-100

-90

-80

-70

P out(d

Bm

)



(b)

1 2 3 4 5 6 7 8 9 10

Frequency (MHz)

-120

-110

-100

-90

-80

-70

P out(d

Bm

)



(c)

Figure 6.14: Impact of using different input power values to the ×6multiplier: (a) RBW=10 kHz, (b) RBW=24 kHz and (c) RBW=39 kHz.


All the noise traces of interest have been measured and results can be observed in Figure 6.14.They show that conversion losses of the multiplier+mixer saturate at about �6 dBm.

6.5 Overall Conclusions from the Measurements

6.5.1 Transmit-receive isolation

The characterization of the ensemble shown in Figure 6.15 has been performed. It producesmore than 60 dB isolation between the transmitter (port 1) and the receiver (port 3) duringthe time in which the transmitter is active. A summary of the obtained results is presented inTable 6.9.

Figure 6.15: Schematic of the isolation chain.

Parameter Measured

Transmit insertion losses (S21) 0.15 dB

Receive insertion losses (S32 ON) 2.2 dB

Receive insertion losses (S32 OFF) 35.6 dB

Transmitter-to-receiver isolation (S31 OFF) 60 dB

Operating band 91 - 98GHz

Table 6.9: Most relevant measurement results of the receiving chain at94GHz.

Transmitter-to-receiver isolation could also be influenced by the amount of reflectionoccurring at the antenna and going through the circulator towards the receiver. However,because the monopulse feed is not yet manufactured, a perfect load was placed at port 2 andthis effect was not accounted for.

72 6.5. OVERALL CONCLUSIONS FROM THE MEASUREMENTS

6.5.2 Maximum output power of the transmitter

Maximum input power rating to avoid permanent damage on the LNA is �20 dBm according tothe datasheet. Therefore, the maximum output power that the transmitter HPA (High PowerAmplifier) should generate must be estimated.

• Power coming from the transmitter might be leaked to the receiver through the circulator:Equation 6.8.

• Power coming from the transmitter that is reflected at the antenna goes through thecirculator again reaching the receiver: Equation 6.9.

PHPA + 20 log(∣∣∣Soff

31

∣∣∣) < −20 dBm (6.8)

PHPA + 20 log(∣∣∣Soff

21

∣∣∣)+ 20 log(∣∣∣SANT

11

∣∣∣)+ 20 log(∣∣∣Soff

32

∣∣∣) < −20 dBm (6.9)

PHPA < min{−20 + 60,−20 + 0.15 + 35.6− SANT

11 (dB)}= min

{40, 15.75− SANT

11 (dB)}(6.10)

Considering that the first pre-designs for the feed feature return losses of 20 - 25 dB, thelimiting factor to the maximum transmitted power will be probably imposed by this feedreflection coefficient[4]. Therefore, until the feed is manufactured and measured, transmitterdesigns should assume an output power no larger than 35 dBm.

6.5.3 Receiver noise floor

The noise figure of the whole receiving chain has been calculated, simulated and measured.Table 6.10 presents a comparison of between the results obtained with those three methods.

An IF active filter has been included at the end of the receiving chain in order to increasethe noise floor of the receiver so that it lies above the floor of the digitalizer. A new active filterwill be designed in the future in order to enlarge the operating band of the system, which shouldhave at least 50MHz. This change will not have much effect on the noise internally generatedby the chain since it is located at the last position.

Receivebandwidth

Pulsewidth

Output noise for Ta=290K

Friis ADS Measurement

39 kHz 25.6 �s �79.74 dBm �77.92 dBm �76.8 dBm

24 kHz 41.7 �s �81.85 dBm �80.03 dBm �78.9 dBm

10 kHz 10 �s �85.65 dBm �83.83 dBm �82.7 dBm

Table 6.10: Comparison of noise power results at the output ofthe receiver chain according to analytical calculations, simulations andmeasurements.

[4]Indeed, the monopulse comparator and the horn for the monopulse-based receiver (Figure 6.1c) are beingdesigned together in order to optimize the isolation between input and output ports of the complete set [16].


6.5.4 Receiver sensitivity

Receiver sensitivity is the minimum received power at the input of the receiver from which thesystem can decide that a target is present. As a reference, in the case of a swerling 5 target, theSNR threshold for detection is 13 dB. In case pulse integration were used, minimum requiredpower would be lower.

Assuming that the noise floor is limited by thermal noise and taking into account previouslytaken noise measurements at the output of the chain, the sensitivity of the receiver can becalculated as Sin = Nout + SNRmin −Grx. Results are presented in Table 6.11.

Receivebandwidth

Minimum power atthe ADC

Receiver sensitivity(minimum power at

the antenna)

39 kHz �63.8 dBm �102.5 dBm

24 kHz �65.9 dBm �104.6 dBm

10 kHz �69.7 dBm �108.4 dBm

Table 6.11: Receiver sensitivity for a swerling 5 target without pulseintegration and Ta = 290K.

74 6.5. OVERALL CONCLUSIONS FROM THE MEASUREMENTS

Chapter 7

Summary and Conclusions

This chapter summarizes the work that has been performed within the realization of this Master’sThesis, along with some of the most important conclusions.

7.1 Summary

This Master’s Thesis has been concerned with the realization of several tasks related to themillimeter-wave subsystem of the space debris radar at 94GHz that is being developed in theframe of the Spaderadar Project [1, 2]. The contributions to this Project that are included inthis document can be summarized in three different tasks:

1. Design of the dual reflector antenna system that will be utilized in millimeter-wave radar.

2. Contribution to the characterization of a reflectarray antenna operating at W-Band.

3. Characterization of the millimeter-wave receiver subsystem of the radar.

The work has been performed in the GMR (Microwave and Radar Group) and the GEA(Applied Electromagnetism Group) of ETSIT-UPM, with punctual collaborations with the GR(Radiation Group).

7.2 Conclusions

Chapter 3 is concerned with the design of the Cassegrain reflector antenna system that will beutilized in the radar. Several simulations have been performed in Grasp yielding a final antennasystem of 900×900×293 mm from which a gain of 57.5 dB is expected. The Cassegrain system isalready being manufactured and, in the future, its performance will be measured in an anechoicchamber and compared to the simulations. Work is currently underway on the final design of amonopulse feed for the antenna system.

Chapter 5 presents the first measurements that have been performed to characterize areflectarray antenna designed to operate at W-Band. As a previous step, the quasi-opticaltest bench for performing free-space measurements had to be designed, for which Chapter 4presents a software based on Gaussian beam propagation theory. This tool was proved to havemany limitations specially because it does not consider reflections and other non-idealities. Theobtained measurement results of the reflectarray show disagreements with respect to what it

75

76 7.2. CONCLUSIONS

was expected from the theoretical model [34]. Even though the phase variation and operationalfrequency range are lower than expected, there are still many measurements that need to beperformed for the purpose of finding the causes of those disagreements and improving the designand manufacturing procedures.

Finally, Chapter 6 involves the characterization of a proposal for the millimeter-wavereceiving subsystem of the radar. On the one hand, the isolation chain has been proved tointroduce about 60 dB isolation, which could effectively protect the receiver if it transmits lessthan 35 dBm. On the other hand, the noise at the output of the receiver chain has been measuredand the introduction of an IF amplifier has been found to be necessary in order to ensure thatthe noise floor of the system is limited by thermal noise. Considering the measurement results,there are still some operative parameters of the radar that have to be selected, such as thetransmitted power or the duration of the pulses. A non-linear characterization of the receivingchain will also have to be performed in the future.

Appendix A

Simulations in Grasp

Grasp is a simulation software from TICRA that uses PO (Physical Optics), PTD (PhysicalTheory of Diffraction) and other numerical methods, to compute scattered fields from surfacesthat are electrically large. This appendix offers a brief introduction on those numerical methods(Section A.1) and details how to configure the software so that a Cassegrain antenna can beaccurately simulated (Section A.2).

A.1 PO and PTD

PO is a numerical method utilized to find the field scattered by a body from the knowledge ofthe incident fields at the part of its surface that is illuminated [19]. PO propagates the fields interms of rays as in an optical method. The following list summarizes the process for obtainingthe total field outside of a scatterer:

• Fields are propagated from source objects towards the scatterers via geometrical optics(ray propagation).

• The induced currents on the surface of scatterers can be determined from the incidentmagnetic field over the surface of the body: �JS = 2n× �Hi

• The scattered field due to the induced currents can be calculated by integrating the currentsover the surface of the object.

• If the scattered field is evaluated asymptotically (farfield), the total field outside the bodycan be described in terms of geometrical optics (rays) by adding it to the incident field.

PO assumes that the scatterers have flat and infinite surfaces. PTD is an extension of POmethod that corrects the PO solution by adding the fields due to the currents close to edges ofthe scatterers [19].

A.2 Grasp configuration for simulating a Cassegrain system

This section details the process to simulate a dual Cassegrain system using Grasp.

77

78 A.2. GRASP CONFIGURATION FOR SIMULATING A CASSEGRAIN SYSTEM

A.2.1 Cassegrain antenna model

The modeling of the dual reflector system is started using the wizard to produce a Cassegrainsystem with default parameters. Then, any object of the system including the feed can bemodified to match the requirements of the simulation.

A.2.2 Command list

Table A.1 illustrates the Grasp commands used for the estimation of the farfield of a Cassegrainantenna system [39].

CommandType

Objects Arguments

1 Get Currents

Field accuracy: -80 dBAuto convergence of PO: on

Source: feed Convergence on scatterer: main reflectorTarget: subreflector Convergence on output grid: farfield cut

Maximum bisections: 5Integration grid limit: on

2 Get Currents


Source: subreflector Convergence on scatterer: subreflectorTarget: main reflector Convergence on output grid: farfield cut


3 Get FieldSource: main reflectorTarget: farfield cut

4 Add FieldSource: feed + subreflectorTarget: farfield cut

5 Get Currents


Source: main reflector Convergence on scatterer:Target: subreflector Convergence on output grid: farfield cut


6 Add FieldSource: subreflectorTarget: farfield cut

Table A.1: Textual reproduction of the Grasp command list.

Simulating the main component of the farfield

The currents induced in the subreflector by fields propagated from the feed (Grasp command 1)produce a scattered field that illuminates the main reflector. This induces other currents in thesurface of the main reflector (Grasp command 2) that scatter to create a pencil-like radiationpattern in broadside direction (Grasp command 3).

APPENDIX A. SIMULATIONS IN GRASP 79

Simulating spillover losses

Spillover and some diffraction effects are considered by adding, to the scattered field from themain reflector, those fields produced at the feed that propagate at an angle close to or greaterthan the semi-subtended angle ΨS , and thus diffract or do not even interact with the reflectors(Grasp command 4).

Simulating blockage from subreflector

Blockage effects have been considered by calculating the currents produced by letting themain reflector scattered field illuminate again the subreflector (Grasp command 5). Thebackwards-propagating field produced by the induced currents is then added to the main reflectorfarfield (Grasp command 6).

80 A.2. GRASP CONFIGURATION FOR SIMULATING A CASSEGRAIN SYSTEM

Appendix B

Simulation Results of DifferentReflection-based Optical TestBenches for 45◦ Incidence

This appendix presents the simulation results that have been obtained for the differentquasi-optical set-ups that were proposed in Section 4.3 to measure the reflection coefficientof a sample impinged at an angle of 45◦ from broadside.

The quasi-optical set-ups are analyzed at 94GHz and at 100GHz and the results arecompared for two different types of horns: Millitech 21 dB horns and RPG 23 dB horns. Theseare the six different afocal quasi-optical systems that are analyzed within the following sections:

• Using two horns and two dielectric lenses: Section B.1.



• Using two horns, two lenses and two 45◦ off-axis mirrors: Section B.4.

• Using two horns, two 45◦ off-axis mirrors and two 90◦ off-axis mirrors: Section B.5



B.1 Optical Set-up Using Two Dielectric Lenses

The optical structures that will be used for this configuration are:

• Two conical horns:

– Millitech smooth wall conical horns with 21 dB in W-band.

– RPG dual mode conical horns with 23 dB in W-band.

• Two plano-convex dielectric lenses of Feq = 62.7mm .

• An ideal flat sample modelled as a plane mirror.

81

82 B.1. OPTICAL SET-UP USING TWO DIELECTRIC LENSES

Figure B.1 shows the schematic of the geometrical arrangement that has been proposed toobtain a beam maximally collimated at the sample position to measure its reflection coefficient.

Input horn

45º

Sample

FbvFfv

Lens

Output horn

Fbv

Ffv

Lens

Figure B.1: Proposed optical arrangementusing horns and lenses.



Lens 1 (convex side) 47.7 0

Sample 131.8 0

Lens 2 (convex side) 131.8 −84.4


Table B.1: Element positions for the proposedarrangement using horns and lensese.

Simulations have been run at 94GHz and 100GHz for the proposed configuration using bothtypes of horns. Simulated results are presented in Figure B.2. In each simulation, the minimumsample diameter that ensures an exp(−2) field decay (�17.4 dB) at its edges has been configured.Obtained minimum sample sizes are presented in Table B.2.

-50 0 50 100 150 200

(mm)

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(a)

-50 0 50 100 150 200

(mm)

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(b)

-50 0 50 100 150 200

(mm)

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(c)

-50 0 50 100 150 200

(mm)

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.2: Simulation results for the sample size that obtains �17.4 dBtaper using two dielectric lenses: (a) 21 dB horns at 94GHz, (b) 21 dB hornsat 100GHz, (c) 23 dB horns at 94GHz and (d) 23 dB horns at 100GHz.

APPENDIX B. OPTICAL TEST BENCHES FOR 45◦ INCIDENCE 83

Parameter21 dB-directivity horns 23 dB-directivity horns

94GHz 100GHz 94GHz 100GHz

Beamwaist (w0) 17.8mm 17.26mm 14.18mm 13.65mm

Sample diameter forexp(−1) field decay

50.5mm 48.9mm 40.25mm 38.8mm


71.5mm 69.3mm 56.95mm 54.9mm

Table B.2: Results at 94GHz and 100GHz obtained using a configurationthat utilizes horns and dielectric lenses for 45◦ incidence.

B.2 Optical Set-up Using Two 45◦ Off-axis Mirrors

The optical structures that will be used in this configuration are:




• Two 45◦ off-axis mirrors:

– 45◦, ∅ = 101.6mm and Feq = 119.03mm.

– 45◦, ∅ = 76.2mm and Feq = 89.28mm (purchase proposal).


B.2.1 Option with available 45◦ off-axis mirrors

The schematic for the proposed optical arrangement to measure the reflection coefficient of thesample using horns and the largest 45◦ off-axis mirrors is presented in Figure B.3.

Paraboloid

mirror

Paraboloid

mirror

Input horn

Output horn

45º

45º

45º

Feq

Feq

FeqSample

Figure B.3: Proposed optical set-up usinghorns and 45◦ off-axis mirrors.



Paraboloid mirror 1 119.03 0

Sample 34.86 −84.17

Paraboloid mirror 2 0 −168.33


Table B.3: Element positions for the proposedset-up using horns and 45◦ off-axis mirrors.

84 B.2. OPTICAL SET-UP USING TWO 45◦ OFF-AXIS MIRRORS

Simulations have been run at 94GHz and 100GHz for the proposed configuration usingboth types of available horns. In each simulation, the minimum sample diameter that ensuresan exp(−2) field decay at its edges has been configured. Results are presented in Figure B.4and Table B.4.

-50 0 50 100 150

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(a)

-50 0 50 100 150

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(b)

-50 0 50 100 150

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(c)

-50 0 50 100 150

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.4: Simulation results for the sample size that obtains �17.4 dBtaper using 45◦ off-axis mirrors with 101.6mm diameter: (a) 21 dB hornsat 94GHz, (b) 21 dB horns at 100GHz, (c) 23 dB horns at 94GHz and (d)23 dB horns at 100GHz.





95.65mm 92.7mm 76.2mm 73.4mm


135.35mm 131.15mm 108mm 103.8mm

Table B.4: Results at 94GHz and 100GHz obtained using a configurationthat uses horns and 45◦ off-axis mirrors with 101.6mm diameter.

B.2.2 Option with alternative 45◦ off-axis mirrors

The purchase of an smaller model of the 45◦ off-axis paraboloid mirrors has been proposed,in order to measure smaller samples. These new mirrors would have an smaller effective focaldistance (89.28mm), which would truncate the beam before it diverges too much. As a downside,the mirrors are smaller: 76.2mm. Results are presented in Figure B.5 and Table B.5.


-50 0 50 100 150

(mm)

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(a)

-50 0 50 100 150

(mm)

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(b)

-50 0 50 100 150

(mm)

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(c)

-50 0 50 100 150

(mm)

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.5: Simulation results for the sample size that obtains �17.4 dBtaper using 45◦ off-axis mirrors with 76.2mm diameter: (a) 21 dB hornsat 94GHz, (b) 21 dB horns at 100GHz, (c) 23 dB horns at 94GHz and (d)23 dB horns at 100GHz.





71.8mm 69.6mm 57.2mm 55.3mm


101.6mm 98.45mm 81mm 77.95mm

Table B.5: Results at 94GHz and 100GHz obtained using a configurationthat uses horns and 45◦ off-axis mirrors with 76.2mm diameter.

B.3 Optical Set-up Using Two 90◦ Off-axis Mirrors





• Two 90◦ off-axis mirrors.

– 90◦, ∅ = 101.6mm and Feq = 152.4mm.


The schematic for the proposed optical arrangement to measure the reflection coefficient ofthe sample using horns and 90◦ off-axis mirrors is presented in Figure B.6. Two different butequivalent structures could be used to obtain the beamwaist at the sample position.

86 B.3. OPTICAL SET-UP USING TWO 90◦ OFF-AXIS MIRRORS

Paraboloid

mirror

Paraboloid

mirror

Input horn

90º

Feq

Feq

Sample

Feq

90º

Output horn

Feq

(a)

Paraboloid

mirror

Paraboloid

mirror

Input horn

90º

45º

Feq

Feq

Sample

Feq

90ºOutput horn

Feq

(b)

Figure B.6: Proposed optical arrangements using horns and 90◦ off-axismirrors: (a) Option 1 and (b) option 2.




Sample 152.4 −152.4

Paraboloid mirror 2 0 −152.4


Table B.6: Element positions for the proposed arrangement (option 2)using horns and 90◦ off-axis mirrors to measure reflectivity at 45◦ incidence.

Simulations have been run at 94GHz and 100GHz for the proposed configuration using bothtypes of available horns. On each simulation, the minimum sample diameter that ensures anexp(−2) field decay at its edge has been configured. Results are presented in Figure B.7 andTable B.7.

-50 0 50 100 150 200

(mm)

-350

-300

-250

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(a)

-50 0 50 100 150 200

(mm)

-350

-300

-250

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(b)

-50 0 50 100 150 200

(mm)

-350

-300

-250

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(c)

-50 0 50 100 150 200

(mm)

-350

-300

-250

-200

-150

-100

-50

0

50

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.7: Simulation results for the sample size that obtains �17.4 dBtaper using 90◦ off-axis mirrors: (a) 21 dB horns at 94GHz, (b) 21 dB hornsat 100GHz, (c) 23 dB horns at 94GHz and (d) 23 dB horns at 100GHz.






122.5mm 118.7mm 97.6mm 93.9mm


173.2mm 167.87mm 138mm 132.85mm

Table B.7: Results at 94GHz and 100GHz obtained using a configurationthat uses horns and 90◦ off-axis mirrors.

Indeed, simulation results show that truncation effects are not limited by the sample size butby the diameter of the mirrors. Therefore, in all the cases, high spillover losses and diffractionwill occur at the edges of the mirror, regardless of the size of the sample.

B.4 Optical Set-up Using Two 45◦ Off-axis Mirrors and TwoDielectric Lenses





• Two 45◦ off-axis paraboloid mirrors:

– 45◦, ∅ = 101.6mm and Feq = 119.03mm.

• Two plano-convex dielectric lenses of Feq = 62.7mm.


Maximum collimation at the sample can be obtained using these optical structures if theyare arranged as indicated in the schematic presented in Figure B.8.

88 B.4. OPTICAL SET-UP USING 45◦ OFF-AXIS MIRRORS AND LENSES

Paraboloid

mirror

Paraboloid

mirror

Input horn

Output horn

45º

45º

45º

Feq

Feq

Sample

FbvFfv

Ffv Fbv

Lens

Lens

Figure B.8: Proposed optical set-up using ofhorns, 45◦ off-axis mirrors and lenses.



Lens 1 (convex side) 47.73 0


Sample 166.66 −84.17

Paraboloid mirror 2 250.82 −168.33

Lens 2 (convex side) 47.73 −168.33


Table B.8: Element positions for the proposedset-up using horns, 45◦ off-axis mirrors and lenses.

Simulations have been run at 94GHz and 100GHz for the proposed configuration using bothtypes of horns. Simulation results are presented in Figure B.2. On each simulation, the minimumsample diameter that ensures an exp(−2) field decay (�17.4 dB) at its edges has been configured.Obtained minimum sample sizes for a successful measurement are presented in Table B.9.

-50 0 50 100 150 200 250 300

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(a)

-50 0 50 100 150 200 250 300

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(b)

-50 0 50 100 150 200 250 300

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(c)

-50 0 50 100 150 200 250 300

(mm)

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.9: Simulation results for the sample size that obtains �17.4 dBtaper using two lenses and two 45◦ off-axis mirrors: (a) 21 dB horns at94GHz, (b) 21 dB horns at 100GHz, (c) 23 dB horns at 94GHz and (d)23 dB horns at 100GHz.






19.5mm 18.9mm 24.3mm 23.75mm


27.8mm 26.95mm 34.65mm 33.8mm

Table B.9: Minimum sample diameter at 94GHz and 100GHz obtainedusing a configuration that utilizes horns and dielectric lenses.

B.5 Optical Set-up Using Two 45◦ Off-axis Mirrors and Two90◦ Off-axis Mirrors


• Two pairs of conical horns:



• Two pairs of off-axis mirrors.

– 45◦, ∅ = 101.6mm and Feq = 119.03mm.

– 90◦, ∅ = 101.6mm and Feq = 152.4mm.


The schematics for the proposed optical arrangements to measure the reflection coefficientof the sample using horns, 45◦ and 90◦ off-axis mirrors are presented in Figure B.10. They arenot equivalent since 45◦ and 90◦ mirrors are not positioned in the same order.

Simulations have been run at 94GHz and 100GHz for the proposed configurations usingboth types of available horns. On each simulation, the minimum sample diameter that ensuresan exp(−2) field decay at its edges has been configured. Results are presented in:

• Figure B.11 and Table B.11 for option 1.

• Figure B.12 and Table B.12 for option 2.

Simulation results show that, in both configurations, truncation effects are not limited bythe sample size, but by the diameter of the 90◦ off-axis mirrors, which is too small. Besides,in the case of option 2, Millitech horns produce an extremely narrow beamwaist at the sample,which doesn’t comply with the paraxial approximation.

90 B.5. OPTICAL SET-UP USING 45◦ AND 90◦ OFF-AXIS MIRRORS

45º

Paraboloid

mirror

45º

45º

F45

90º

Paraboloid

mirror

90º

Sample

Input horn

45º

F45

90º

Paraboloid

mirror

90º

45º

Paraboloid

mirror

(a)

45º

Paraboloid

mirror

Input horn

45º

F90 + F45Sample

F90

90º

45º

90º

Paraboloid

mirror

45º

Paraboloid

mirror

Output horn

45º

F90 + F45

F90

90º

90º

Paraboloid

mirror

(b)

Figure B.10: Proposed optical arrangements using of horns, 45◦ and 90◦

off-axis mirrors: (a) Option 1 and (b) option b.

Optical elementOption 1 Option 2

x(mm) y(mm) x(mm) y(mm)

Horn 1 (phase center) 0 0 0 0

45◦ off-axis mirror 1 119.03 0 152.4 −271.43

90◦ off-axis mirror 1 −72.9 −191.93 152.4 0

Sample 180.7 84.17 236.57 −187.26

45◦ off-axis mirror 2 −288.4 −191.9 320.73 0

90◦ off-axis mirror 2 96.5 −383.9 320.73 −271.43

Horn 2 (phase center) 215.5 383.9 473.13 0

Table B.10: Element positions for the proposed arrangement using horns,45◦ and 90◦ off-axis mirrors.

B.5.1 Results for option 1





13.3mm 12.9mm 16.6mm 16.2mm


19.5mm 18.7mm 23.8mm 23.2mm

Table B.11: Results at 94GHz and 100GHz obtained using aconfiguration that uses horns, 45◦ and 90◦ off-axis mirrors (option 1).


-300 -250 -200 -150 -100 -50 0 50 100 150 200

(mm)

-500

-450

-400

-350

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(a)

-300 -250 -200 -150 -100 -50 0 50 100 150 200

(mm)

-500

-450

-400

-350

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(b)

-300 -250 -200 -150 -100 -50 0 50 100 150 200

(mm)

-500

-450

-400

-350

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(c)

-300 -250 -200 -150 -100 -50 0 50 100 150 200

(mm)

-500

-450

-400

-350

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.11: Simulation results for the sample size that obtains �17.4 dBtaper using 45◦ and 90◦ off-axis mirrors (option 1): (a) 21 dB horns at94GHz, (b) 21 dB horns at 100GHz, (c) 23 dB horns at 94GHz and (d)23 dB horns at 100GHz.

92 B.5. OPTICAL SET-UP USING 45◦ AND 90◦ OFF-AXIS MIRRORS

B.5.2 Results for option 2

-100 -50 0 50 100 150 200 250 300 350 400 450 500

(mm)

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(a)

-100 -50 0 50 100 150 200 250 300 350 400 450 500

(mm)

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(b)

-100 -50 0 50 100 150 200 250 300 350 400 450 500

(mm)

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(c)

-100 -50 0 50 100 150 200 250 300 350 400 450 500

(mm)

-300

-250

-200

-150

-100

-50

0

50

100

(mm

)


Beam e-1

Beam e-2

(d)

Figure B.12: Simulation results for the sample size that obtains �17.4 dBtaper using 45◦ and 90◦ off-axis mirrors (option 2): (a) 21 dB horns at94GHz, (b) 21 dB horns at 100GHz, (c) 23 dB horns at 94GHz and (d)23 dB horns at 100GHz.





9.5mm 8.2mm 10.4mm 10.1mm


13.1mm 12.55mm 15.4mm 14.95mm

Table B.12: Results at 94GHz and 100GHz obtained using aconfiguration that uses horns, 45◦ and 90◦ off-axis mirrors (option 2).

Appendix C

Phase Center Calculation

The phase center location of the horns utilized as feeds has to be known ir order to place themat their optimum positions. Even though there are some analytical approaches to find the phasecenter location of a horn [10], in this case, a numerical approach based on simulations has beenpreferred. HFSS, which is a simulator based on FEM (Finite Element Method), has been used.The procedure that has been followed can be summarized in the following four steps:

• Creation of the HFSS model of the antenna.

• Definition of a relative coordinate system that is offset from the global coordinate systemin z-dimension. Its origin must be located at a certain distance pos from the aperture ofthe horn (if pos > 0, then the origin of the relative coordinate system is inside the horn).

• Creation of a rectangular plot for the phase of the main cuts of the E-field copolarcomponent multiplied by the distance R.

• Modification of the position of the relative coordinate system until the flattest possibleresponse is obtained in the phase plot. A parametric study is also possible.

C.1 RPG FH-PP-100 Potter Horn

The phase center of the RPG FH-PP-100 conical horn is calculated in this section. The datasheetof this horn, which includes its dimensions, can be found in Appendix G.

Figure C.1 presents the HFSS model that has been used to simulate the horn. Takingadvantage of the modal and geometrical symmetries of the antenna, it was only necessary tosimulate a quarter of the model. Magnetic and electric symmetries must be applied carefully atthe corresponding splitting planes.

93

94 C.1. RPG FH-PP-100 POTTER HORN

Figure C.1: HFSS model of the RPG horn using symmetries.

After several iterations looking for the coordinate system origin for which the phase of theelectric field is maximally planar, the phase center of the horn has been found. The horn responsehas been studied at 94GHz and 100GHz, since these are the center frequencies at which thehorn will be used. A summary is presented in Table C.1. Phases at 94GHz and 100GHz arepresented in Figure C.2 and Figure C.3 and directivities in Figure C.4 and Figure C.5.

FrequencyPhase center

positionMaximum variationin a 16◦-beamwidth

Directivity

94GHz 20.5mm 9.81◦ 23.14 dB

100GHz 24.5mm 13.29◦ 23.43 dB

Table C.1: Phase center at different frequencies measured from theaperture (z > 0 is inside the horn).

Figure C.2: Phase of the copolar component of the electric field at 94GHzwhen the coordinate system is at z = 20.5mm inside the horn.

APPENDIX C. PHASE CENTER CALCULATION 95

Figure C.3: Phase of the copolar component of the electric field at100GHz when the coordinate system is at z = 24.5mm inside the horn.

Figure C.4: Directivity at 94GHz.


96 C.2. MILLITECH SGH-08 CONICAL HORN

C.2 Millitech SGH-08 Conical Horn

The phase center of a Millitech SGH-08 conical horn is calculated in this section. The datasheetof this horn, which includes its dimensions, can be found in Appendix G.

Figure C.6 presents the HFSS model that has been used to simulate this horn. Takingadvantage of the modal and geometrical symmetries only a quarter of the antenna is simulated.Magnetic and electric symmetries must be applied carefully at the corresponding splitting planes.

Figure C.6: HFSS model of the Millitech horn using symmetries.

After several iterations looking for the coordinate system origin for which the phase of theelectric field is maximally planar, the phase center of the horn has been found. The horn responsehas been studied at 94GHz and 100GHz, since these are the center frequencies at which thehorn will be used. A summary is presented in Table C.2. Phases at 94GHz and 100GHz arepresented in Figure C.7 and Figure C.8 and directivities in Figure C.9 and Figure C.10.

FrequencyPhase center

positionMaximum variationin a 20◦-beamwidth

Directivity

94GHz 6.6mm 4.75◦ 20.67 dB

100GHz 7.9mm 6.17◦ 21.08 dB

Table C.2: Phase center at different frequencies measured from theaperture (z > 0 is inside the horn).

APPENDIX C. PHASE CENTER CALCULATION 97

Figure C.7: Phase of the copolar component of the electric field at 94GHzwhen the coordinate system is at z = 6.6mm inside the horn.

Figure C.8: Phase of the copolar component of the electric field at100GHz when the coordinate system is at z = 7.9mm inside the horn.


98 C.2. MILLITECH SGH-08 CONICAL HORN


Appendix D

Measurement Set-ups andEquipment Configurations

This appendix summarizes the measurement equipment, set-ups and configurations thathave been used along the thesis to characterize some of the components belonging to themillimeter-wave subsystem of the radar.

D.1 S-Parameters Characterization of W-band Devices

An Anritsu broadband VNA has been used for the characterization of S-parameters of activeand passive devices at W-Band (75 - 110GHz). All the equipment used for this measurement issummarized below:

• An Anritsu ME7838B broadband VNA operational from 70 kHz to 110GHz. It consistsof the following items:

– A MS4647B VectorStar VNA, 70 kHz to 70GHz.

– Two 3743A millimeter-wave modules to extend the band up to 110GHz.

– A 3739B broadband millimeter-wave test set and interface cables.

• SAGE Millimeter SWC-101F-E1 1mm-coaxial (F) to WR-10 waveguide transitions.

• TRL (Through-Reflect-Line) calibration kit in WR-10.

• WR-10 waveguide terminations to match the ports.

• WR-10 straight pieces of waveguide to facilitate connections.

The measurement set-up for any of the measurements is shown in Figure D.1. Basically,the DUT (Device Under Test) must be connected to the output ports of the VNA. A shortstraight waveguide is collocated between the DUT and port 2 in order to facilitate connections(see Figure D.1b).

99

100 D.1. S-PARAMETERS CHARACTERIZATION OF W-BAND DEVICES

VNA display

Test Set

VNA

control

mm-Wave

moduleDUT

mm-Wave

moduleDUTDUT

Coax to WR-10

(a) (b)

Figure D.1: Measurement set-up used for the acquisition of theS-parameters of millimeter-wave devices.

Table D.1 summarizes the VNA configuration for which the calibration and the S-parametersmeasurements are performed.

Parameter Value

Sweep mode Frequency sweep

Frequency range 75 - 110GHz

RF Power�30 dBm

IF bandwidth 10 kHz

Number of points 3201 points

Calibration Manual TRL

Trace averaging None

Table D.1: Anritsu VNA configuration for the measurement ofS-parameters at W-band.

Calibrating the VNA in frequency allows compensation of all those time-invariantmeasurement errors of the system.

Finally, *.s2p files are post-processed using Matlab. The whole measuring process has thefollowing steps:

1. Calibration of the VNA.

2. S-Parameters acquisition in the whole calibrated band (75 - 110GHz).

3. Correction of measurement imperfections.

4. Application of a smoothing filter to the traces in order to eliminate residual noise. Thisfiltering consists of a 2% averaging window.

5. Zooming into the band of interest of 88 - 102GHz. For this particular radar applicationthe band of interest is even narrower, being 92 - 96GHz.

APPENDIX D. MEASUREMENT SET-UPS AND EQUIPMENT CONFIGURATIONS 101

AW-band load is needed to match ports that are not being measured during the S-parameterscharacterization of devices with more than 2 ports, and also as a noise source to produce noiseat W-band. A Quinstar W-band load has been used for those two purposes.

Obtained results (Figure D.2) are consistent with the expected VSWR (Voltage StandingWave Ratio) from the datasheet and prove the usefulness of the load as a 50Ω “perfect” resistiveload.

75 80 85 90 95 100 105 110

Frequency (GHz)

-80

-70

-60

-50

-40

-30

-20

-10

0

10

(dB

)

S-Parameter

S11

(a)

75 80 85 90 95 100 105 110

Frequency (GHz)

1

1.02

1.04

1.06

1.08

1.1

1.12

1.14

1.16

VWSR

VSWR

MeasuredDatasheet

(b)

Figure D.2: Measurement of a Quinstar W-band load: (a) Reflectioncoefficient and (b) VSWR.

D.2 Noise Measurements

Since a W-band noise source is not currently available in the laboratory, noise at W-band isgenerated using the W-band load from Figure D.2. Noise performance of the whole receivingchain has been characterized by obtaining the noise power level at the intermediate frequency(after downconversion by the mixer) using a microwave spectrum analyzer. All the equipmentused for this measurement is summarized below:

• Signal generator from Keysight.

• An Agilent EXA N9010 spectrum analyzer.

• WR-10 waveguide terminations to match the ports.

• WR-10 straight pieces of waveguide to facilitate connections.

Basically, loads of 50Ω are used to match those ports of the millimeter-wave subsystem thatare not connected to the receiver. These (ideally) perfect matched loads would act as noisesources with a noise temperature of T = 290K. Afterwards, the obtained response will becompared to the expected values from Friis formula and ADS noise budget simulations.

The measurement set-up for any of the noise measurements that have been performed ispresented in Figure D.3, and Table D.2 summarizes the spectrum analyzer configuration forwhich the noise measurements are performed.

102 D.3. FREE-SPACE MEASUREMENTS

IF componentsmm-wave

components

mixerZ0

IFRF

Signal

generator

Spectrum analyzer

Figure D.3: Measurement set-up for the acquisition of the noise powerlevel at the output of the receiving chain

Parameter Value


Frequency span 0 - 10MHz

RBW (Resolution Bandwidth) FFT bin width

VBW (Video Bandwidth) 1000

Attenuation 0 dB

Trace averaging 100 passes

Detector type Average RMS

Table D.2: Agilent spectrum analyzer configuration for the measurementof noise level of the receiving chain.

Obtained measurements will be post-processed using Matlab. The whole noisecharacterization process has the following steps:

1. Configuration of the spectrum analyzer, so that the noise floor level is at least 10 dB belowthe measurements.

2. Noise power level acquisition.

3. Correction of measurement imperfections.

4. Application of a smoothing filter to the traces in order to eliminate residual noise.

5. Zooming into de band of interest and comparison of the measured results with thecalculated and simulated results.

D.3 Free-Space Measurements

In Section 5.3, different optical configurations have been designed to measure the reflectioncoefficient of a reflectarray sample impinged from different angles. In order to capture theresponse of the reflectarray sample, the output of the horns has been connected to an Anritsubroadband VNA.

APPENDIX D. MEASUREMENT SET-UPS AND EQUIPMENT CONFIGURATIONS 103

The measurement set-up is similar to what has been used in Section D.1. Although, in thiscase, the DUT of Figure D.1 would be the entire quasi-optical system. 1mm-long coaxial cablesare connected between the horns and the output ports of VNA in order to facilitate connections.This condition can be observed in Figure D.4.

Figure D.4: Coaxial cable and WR-10 transitions between the horns andthe VNA ports.

Table D.3 summarizes the VNA configuration for which the calibration and the S-parametermeasurements are performed.

Parameter Value

Sweep mode Frequency sweep

Frequency range 80 - 110GHz

RF Power�30 dBm

IF bandwidth 10 kHz


Calibration Manual TRL

Trace averaging None

Table D.3: Anritsu VNA configuration for the free-space measurementsat W-band.

Calibration of the VNA is performed at the input port of the horns, due to the lack ofa standard procedure to calibrate a reflection-based quasi-optical system. This compensatesthe time-invariant measurement errors, that occur up to the input waveguide of the horns.Calibration accuracy is limited to about 25 dB, since coaxial cables had to be moved during thecalibration to connect the thru. This can be observed in Figure D.5.

104 D.3. FREE-SPACE MEASUREMENTS

80 85 90 95 100 105 110

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

Calibration

S11S12S21S22

Figure D.5: Thru of the TRL calibration kit after calibration.

Post-processing is performed using Matlab and it is explained in detail in Appendix E. Thewhole measurement process can be summarized in the following steps:

1. Calibration of the VNA.

2. S-parameters acquisition of the thru.

3. S-Parameters acquisition of the sample under test.

4. Re-calibration of the measured S21 of the sample with the measured S21 of the thru.

5. Correction of measurement imperfections:

– Time domain processing for the removal of uncalibrated reflections.

– Application of a smoothing filter to the traces in order to eliminate residual noise.

Appendix E

Free-space Post-processingTechniques

This appendix summarizes the post-processing techniques that have been utilized to correct themeasurement errors, reflections and residual noise that are present on the free-space acquisitions.

E.1 Recalibration with a Reference

As commented in Appendix D, calibration of the measurement system has been performed at theinput waveguide of the horns. Therefore, reflections and losses occurring inside the quasi-opticalbench are not being compensated.

Three measurements would be necessary for a complete free-space calibration. But due to thelack of three accurate calibration standards for this set-up, a single optical thru has been utilizedas reference sample to compensate the losses and part of the ripple associated to quasi-opticalelements. This “optical thru” consists of a flat metallic surface which has the same size as thesample under test. A photograph of the measurement set-up for acquiring the response of thethru is presented in Figure E.1.

Figure E.1: Photograph of the set-up to measure the optical thru.

The response of the reflectarray is embedded in the transmission coefficient of the measuredS-parameters. Therefore, the losses and phase shifts introduced by the optical system can becoarsely compensated using Equation E.1, which yields the results observed in Figure E.2.

105

106 E.2. TIME-DOMAIN ANALYSIS

S21 corr =S21 sample

S21 thru(E.1)

The measured reflectarray response will always lie below the losses of the reference thru. Onthe other hand, if any resonance peak falls below the calibration precision in a certain band, thecurve will be corrupted with noise in that band. Therefore, in order to maximize the dynamicmargin of the measurement system, the optical test bench should introduce the lowest possiblelosses (Sthru

21 → 0 dB) and the calibration level must as low as possible.

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-30

-25

-20

-15

-10

-5

0

(dB

)

Re-calibrated amplitude of the transmission coefficient

SampleThruCalibrated sample

(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-150

-100

-50

0

50

100

150

(dB

)

Re-calibrated angle of the transmission coefficient

SampleThruCalibrated sample

(b)

Figure E.2: Calibration of the transmission coefficient with an opticalthru: (a) Amplitude and (b) phase.

E.2 Time-domain Analysis

As it can be observed in Figure E.2, ripple associated to reflections occurring inside themeasurement system is not completely eliminated by Equation E.1.

A time domain analysis has been performed, for the purpose of observing the most powerfulreflections of the system. Time domain responses obtained using a rectangular window arepresented in Figure E.3:

• Sample response zone is marked in both Figure E.3a and Figure E.3b. It is approximatelycoincident for S11 and S21 due to the symmetry of the optical configuration. That zonewould be the gated-in band after time-domain processing.

• The positions of the identified discontinuities are consistent with their real position in themeasurement system.

• The reflectarray introduces a delay with respect to the thru. This is because the reflectarrayis formed by several layers (with different and tunable permittivities) and the thru isaligned with the quartz superstate (see Figure 5.2).

APPENDIX E. FREE-SPACE POST-PROCESSING TECHNIQUES 107

0 500 1000 1500 2000 2500 3000

Time (ps)

0

10

20

30

40

50(d

B)

Reflection coefficient in time domain

S11-ThruS11-Sample

Input face of lens 2

Output face of lens 2

Output face of lens 2Input face of lens 1

Aperture of the horn

Sampleresponse

Input

horn

30º

Sample

FbvFfv

Lens

Output

horn

Lens

(a)

0 500 1000 1500 2000 2500 3000

Time (ps)

0

10

20

30

40

50

60

(dB

)

Transmission coefficient in time domain

S21-ThruS21-Sample

Thru

LC layer

Sampleresponse

Ground planeDipoles layer

Superstrate layer

Input

horn

30º

Sample

FbvFfv

Lens

Output

horn

Lens

(b)

Figure E.3: Time domain transformation of the S-parameters of the30◦-incidence optical test bench: (a) S11 and (b) S21.

E.3 Smoothing vs Time-domain Processing

Two different procedures are proposed to eliminate the uncalibrated reflections and residualerrors:

• Smoothing filter:

1. Calibration of the sample measurement with the thru.

2. Application of a smoothing filter to the corrected transmission coefficient. This filterwould consist of an averaging window of length 5% of the samples.

• Time domain processing:

1. Filtering of the time-domain response of the thru measurement.

2. Filtering of the time-domain response of the sample measurement.

3. Transformation of time-gated responses of the thru and the sample back to frequencydomain.

108 E.3. SMOOTHING VS TIME-DOMAIN PROCESSING

4. Calibration of the time-gated sample with the time-gated thru.

Results from both methods are compared in Figure E.4. It can be concluded that they bothyield very similar responses and both reduce the ripple present on the measured S21 curve.

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-25

-20

-15

-10

-5

0

(dB

)

Post-processed amplitude of the transmission coefficient

CalibratedSmoothedTime-filtered

(a)

88 90 92 94 96 98 100 102 104 106

Frequency (GHz)

-150

-100

-50

0

50

100

150

(dB

)

Post-processed phase of the transmission coefficient

CalibratedSmoothedTime-filtered

(b)

Figure E.4: Post-processing of the transmission coefficient: (a) Amplitudeand (b) phase.

Appendix F

Characterization of Millimeter-waveComponents

This appendix presents the detailed characterization of some of the most relevant componentsthat will be used in the radar receiving chain, if their response is considered appropriate for theapplication:

• Passive devices:

– RPG WFI-110 isolator.

– RPG WPD-110 hybrid power divider.

– ELVA CR-1094 circulator.

– Quinstar QAL-W00000 variable attenuator.

– ELVA SPST-10 switch.

• Active devices:

– RPG W-LNA75110 low noise amplifier.

In particular, the S-parameters of each specific component have been measured:

• Measurements and conclusions can be found in the sections associated to each component.

• The necessary equipment and its specific configuration can be found in Appendix D.

• The manufacturers’ datasheets corresponding to each millimeter-wave component can befound in Appendix G.

109

110 F.1. RPG WFI-110 ISOLATOR

F.1 RPG WFI-110 Isolator

This RPG isolator will be used at the input of the reflective switch to attenuate its reflectionswhen it is switched-off.

• Figure F.1 shows the measurement set-up that has been used for its characterization.

• Measured S-parameters are presented in Figure F.2.

• Insertion losses, return losses and isolation and group delay can be observed in detail inFigure F.3.

Figure F.1: Measurement set-up for the RPG WFI-110 isolator.

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-Parameters

S11 measuredS12 measuredS21 measuredS22 measuredRL datasheetIS datasheetIL datasheet

(a)

85 90 95 100 105

Frequency (GHz)

-150

-100

-50

0

50

100

150

(º)

S-Parameters (phase)

S11S21

(b)

Figure F.2: RPG WFI-110 isolator: (a) S-Parameters amplitude and (b)S-Parameters phase.

As shown in Figure F.3a, this device presents insertion losses in the band of interest(92 - 96GHz) of 1.6 dB, which are 0.3 dB above the manufacturer specification. Isolation(Figure F.3c) is higher than 20 dB across the whole W-band. Additionally Figure F.3b showsreturn losses better than 16.5 dB in the band of interest. Group delay presents a flat response(Figure F.3d) across the whole band of interest, which is positive in case of using widebandsignals.

Table F.1 presents a summary of the measurement results that have been obtained for thisdevice.

APPENDIX F. CHARACTERIZATION OF MILLIMETER-WAVE COMPONENTS 111

90 91 92 93 94 95 96 97 98

Frequency (GHz)

1

1.2

1.4

1.6

1.8

2

IL (d

B)

Insertion Losses

MeasuredDatasheet

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

15

20

25

30

35

RL

(dB

)

Return Losses

Input PortOutput PortDatasheet

(b)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

15

20

25

30

Isol

atio

n (d

B)

Isolation

MeasuredDatasheet

(c)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

0.1

0.2

0.3

0.4

0.5

g(ns)

Group delay

Measured

(d)

Figure F.3: RPG WFI-110 isolator: (a) Insertion losses, (b) return losses,(c) isolation and (d) group delay.

Parameter Measured Datasheet

Insertion Losses 1.6 dB 1.3 dB

Return losses 19.7 dB 14 dB


Table F.1: RPG WFI-110 isolator measurement results at 94GHz incomparison with those provided by the manufacturer.

F.2 RPG WPD-110 Hybrid Power Divider

This hybrid power divider might be used to divide the LO signal, so that it can be used todown-convert the different channels of the monopulse architecture.

Figure F.4 shows the measurement set-up that has been used for the characterization ofthis RPG power divider. As observed, a straight piece of rectangular waveguide is strictlynecessary, in order to collocate the load at the corresponding port. The load used for matchingis characterized in Appendix D and it can be thought as an ideal Z0.

• Measured S-parameters are presented in Figure F.5. It was not possible to measure the

112 F.2. RPG WPD-110 HYBRID POWER DIVIDER

P2

P3

P1

Figure F.4: Measurement set-up for the RPG WPD-110 power divider.

isolation between ports 2 and 3 because of their physical closeness (see Figure F.4).

• Insertion and return losses are shown with more detail in Figure F.6.

• Relative phase shift between both paths and group delays are presented in Figure F.7.

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-parameters

S11S12S13S21S22S31S33

(a)

85 90 95 100 105Frequency (GHz)

-150

-100

-50

0

50

100

150

(º)

S-parameters (phase)

S21S31

(b)

Figure F.5: RPG WPD-110 hybrid power divider: (a) S-Parametersamplitude and (b) S-Parameters phase.

Insertion losses (Figure F.6a) of the power divider are slightly higher at 94GHz than thoseprovided by the manufacturer. Amplitude balance between both ports is in agreement with theexpected hybrid behaviour, since the difference ranges from 0.3 to 0.8 dB in the band of interest(92 - 96GHz).

In addition, input return losses (Figure F.6b) are better than 13 dB across the whole band ofinterest, limiting on the lower part of this band (92GHz). In general their value is much higherthan what is indicated by the manufacturer.


90 91 92 93 94 95 96 97 98

Frequency (GHz)

2.5

3

3.5

4

4.5

5

5.5

IL (d

B)

Insertion Losses

S21S31Datasheet S21Datasheet S31

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

5

10

15

20

25

30

RL

(dB

)

Return Losses

Port 1Port 2Port 3Datasheet

(b)

Figure F.6: RPG WPD-110 hybrid power divider: (a) Insertion lossesand (b) return losses.

The phase responses of S21 and S31, shown in Figure F.7a, present a deviation of 180± 10◦

in the band of interest (92 - 96GHz). Therefore the divider is designed as a 180◦-hybrid, whichis not indicated in the datasheet.

Calculated group delay (Figure F.7b) is very small and quite flat across the whole band andfor both traces. Besides, group delay balance ensures that two signals coming from differentports would suffer more or less the same delay.

90 91 92 93 94 95 96 97 98Frequency (GHz)

160

165

170

175

180

185

190

195

200

|(S

21)-

(S31

)|(º)

Phase balance

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

50

100

150

200

250

g(ps)

Group delay

S21S31

(b)

Figure F.7: RPG WPD-110 hybrid power divider: (a) phase balance and(b) group delay.

Table F.2 summarizes the measurement results obtained for this device at 94GHz.

114 F.3. ELVA CR-1094 CIRCULATOR


Insertion Losses (1 → 2) 3.28 dB 3 dB

Coupling Losses (1 → 3) 3.81 dB 3.6 dB

Return losses > 16.1 dB 6.5 dB

Phase balance 176◦ —

Table F.2: RPG WPD-110 hybrid power divider measurement results at94GHz in comparison with those provided by the manufacturer.

F.3 ELVA CR-1094 Circulator

This ELVA narrow band circulator is meant to isolate the transmit and receive subsystems. Thecirculator response has been obtained connecting a pair of its ports to the VNA and matchingthe third port with a waveguide termination which is characterized in Appendix D. Threemeasurements are performed to cover any possible combination of ports. Measurement set-upscan be observed in Figure F.8.

2

1

3

(a)

2

1

3

(b)

2

13

(c)

Figure F.8: Measurement set-up for the ELVA CR-1094 circulator: (a)ports 1-2, (b) ports 1-3 and (c) ports 2-3.

Measured S-parameters are presented in Figure F.9 before and after load correction [1]. Asobserved, there are no major changes in the curves after load correction(Figure F.9b), which iscoherent with the fact that an almost perfect load (ρL ≈ 0) has been used to match the unusedport at each measurement.

[1]Load correction refers to a correction technique that compensates the use of imperfect loads to match theunused ports during a two-port measurement of a device with more than two ports.


85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-parameters

S11S12S13S21S22S23S31S32S33

(a)

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-parameters

S11S12S13S21S22S23S31S32S33

(b)

Figure F.9: ELVA CR-1094 circulator: (a) Measured S-Parametersamplitude (referred to the real load) and (b) S-Parameters amplitude afterload correction (referred to Z0).

Insertion losses, return losses and isolation are shown in Figure F.10. This device presentsvery low insertion losses (Figure F.10a) of only 0.3 dB and return losses (Figure F.10b) betterthan 26.5 dB, which are values above the specification. In addition, it features an isolation(Figure F.10c) better than 22 dB across the whole band of interest. In general, it can be observedthat the band in which the circulator works correctly ranges from 91 to 98GHz.

Table F.3 summarizes the measurement results obtained for this device.


Insertion Losses 0.22 dB 0.7 dB

Return Losses 26.7 dB 20.8 dB

Isolation 25 dB 23 dB

Bandwidth 7GHz 2.5GHz

Table F.3: ELVA CR-1094 circulator measurement results at 94GHz incomparison with those provided by the manufacturer.

116 F.4. QUINSTAR QAL-W00000 VARIABLE ATTENUATOR

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

0.2

0.4

0.6

0.8

1

IL(d

B)

Insertion Losses

S21S32S13Datasheet

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

15

20

25

30

35

40

45

50

RL

(dB

)

Return Losses

Port 1Port 2Port 3Datasheet

(b)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

15

20

25

30

35

40

45

Isol

atio

n (d

B)

Isolation

S12S23S31Datasheet

(c)

Figure F.10: ELVA CR-1094 circulator: (a) Insertion losses, (b) returnlosses and (c) isolation.

F.4 Quinstar QAL-W00000 Variable Attenuator

Figure F.11 shows the measurement set-up that has been used for the characterizations of thisQuinstar variable attenuator.

Figure F.11: Measurement set-up for the Quinstar QAL-W00000 variableattenuator.

Different measurements have been taken to compare the response of different attenuationpositions. Obtained results for different positions are presented in Figure F.12 and Figure F.13.


90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

15

20

25

30

35

40

45

RL

(dB

)Input Return Losses

Position 0Position 5Position 10Position 15Position 20Position 25Datasheet (max)

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

15

20

25

30

35

40

45

RL

(dB

)

Output Return Losses


(b)

Figure F.12: Quinstar QAL-W00000 attenuator: (a) Input return lossesand (b) output return losses

Input and output return losses (Figure F.12a and Figure F.12b, respectively) are above 17 dBacross the whole W-band, which is in agreement with the expected values extracted from thedatasheet.

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

5

10

15

20

25

Att

(dB

)

Attenuation


(a)

0 5 10 15 20 25 30

Micrometer position

0

5

10

15

20

Att

(dB

)

Attenuation

92-96 GHz

(b)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

-150

-100

-50

0

50

100

150

S21

(º)

Phase Shift

Position 0Position 5Position 10Position 15Position 20Position 25

(c)

Figure F.13: Quinstar QAL-W00000 attenuator: (a) Attenuation vs.frequency, (b) attenuation vs. micrometer position and (c) phase shift.

118 F.5. ELVA SPST-10 SWITCH

Figure F.13a shows that the maximal attenuation that can be obtained with this device is20 dB (position 0), which is 5 dB lower than the manufacturer specification. In addition, it canbe observed (Figure F.13a) that the micrometer position is not linearly related to the obtainedattenuation. Figure F.13b presents an approximate correspondence between each micrometerposition and the mean attenuation level that would be obtained for that position in the band ofinterest (92 to 96GHz).



Return Losses 23 dB 14 dB

Maximum attenuation 20 dB 25 dB

Table F.4: Quinstar QAL-W00000 attenuator measurement results at94GHz in comparison with those provided by the manufacturer.

F.5 ELVA SPST-10 Switch

This ELVA switch is used in reception, just after the circulator, in order to provide higherisolation to the active components of the receiver during transmitting periods. It will be in theoff-state during transmission periods and in the on-state during reception periods. Figure F.14shows the measurement set-up that has been used for its characterization.

Figure F.14: Measurement set-up for the ELVA SPST-10 switch.

The necessary biasing conditions to obtain off- and on-states are presented in Table F.5.

StateReferencevoltage

Positivesupply (5V)

current

Negativesupply (�9V)

current

Total powerconsumption

OFF 0V 34mA 29mA 0.43W

ON 5V 100mA 20mA 0.68W

Table F.5: ELVA SPST-10 switch: biasing and power consumption.

• Measured S-parameters are shown in Figure F.15.


• Insertion losses for each state can be seen in detail in Figure F.16.

• Return losses are presented in Figure F.17.

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (OFF)

S11S12S21S22

(a)

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (ON)

S11S12S21S22

(b)

Figure F.15: ELVA SPST-10 switch: (a) S-Parameters amplitude whenthe switch is OFF and (b) S-Parameters amplitude when the switch is ON.

This switch shows insertion losses (Figure F.16b) lower than 1.1 dB in the band of interest(92 to 96GHz) when it is biased in the closed state (reference voltage level of 5V).

When the switch is off-biased (reference voltage level of 0V) an isolation larger than 31 dB isobtained for the whole band of interest (Figure F.16a). This high level of isolation will protectthe receiver from leakage of transmit power occurring during transmission periods.

As observed in Figure F.17a, this is a reflective switch, since most of the RF power that isnot being transmitted in the off-state is actually being reflected. Its extremely low return lossesshow the necessity of an isolator at the input of the switch, to prevent reflected power frombeing re-radiated. (see Section F.6).

90 91 92 93 94 95 96 97 98

Frequency (GHz)

25

30

35

40

45

IL (d

B)

Insertion Losses (OFF)

S21(OFF)S12(OFF)Datasheet (typ)

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0.5

1

1.5

IL (d

B)

Insertion Losses (ON)

S21(ON)S12(ON)Datasheet (typ)

(b)

Figure F.16: ELVA SPST-10 switch: (a) Insertion losses when the switchis OFF and (b) insertion losses when the switch is ON.


120 F.6. JOINT RESPONSE OF THE ISOLATOR AND THE SWITCH

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

0.2

0.4

0.6

0.8

1

RL

(dB

)Return Losses (OFF)

S11(OFF)S22(OFF)

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

10

15

20

25

30

RL

(dB

)

Return Losses (ON)

S11(ON)S22(ON)

(b)

Figure F.17: ELVA SPST-10 switch: (a) Return losses when the switchis OFF and (b) return losses when the switch is ON.


Insertion losses (S21 ON) 0.96 dB 1 dB

Isolation (S21 OFF) 35.1 dB > 30 dB

Return loss (S11 OFF) 0.55 dB —

Table F.6: ELVA SPST-10 switch measurement results at 94GHz incomparison with those provided by the manufacturer.

F.6 Joint Response of the Isolator and the Switch

The extremelly low return losses of the off-biased switch (see Section F.5) demand an isolatorat its input to attenuate reflected power. Therefore, this ensemble is expected to achieve:

• Improved return losses during the off-state thanks to the attenuation of the reflected powerfrom the switch in the isolator.

• Low insertion losses in the on-state because of the low losses provided by both the switchand the isolator.

The joint response of both devices will be presented in this section for both statesof the switch. Figure F.18 shows the measurement set-up that has been utilized for thecharacterization.


Isolator

Switch

Figure F.18: Measurement set-up to characterize the joint response ofthe isolator and the switch.

• Measured S-parameters are shown in Figure F.19.

• Return losses are presented in detail for each state Figure F.20.

• Insertion losses for each state are presented in detail in Figure F.21.

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (OFF)

S11S12S21S22

(a)

85 90 95 100 105

Frequency (GHz)

-60

-50

-40

-30

-20

-10

0

(dB

)

S-parameters (ON)

S11S12S21S22

(b)

Figure F.19: Isolator + switch: (a) S-Parameters amplitude when theswitch is OFF and (b) S-Parameters amplitude when the switch is ON.

122 F.6. JOINT RESPONSE OF THE ISOLATOR AND THE SWITCH

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

5

10

15

20

25

30

RL

(dB

)

Input Return Losses (OFF)

Isolator+SwitchSwitchIsolator

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

14

16

18

20

22

24

26

28

30

RL

(dB

)

Input Return Losses (ON)


(b)

Figure F.20: Isolator + switch: (a) Return losses when the switch is OFFand (b) return losses when the switch is ON.

90 91 92 93 94 95 96 97 98

Frequency (GHz)

30

32

34

36

38

40

42

IL (d

B)

Insertion Losses (OFF)

Isolator+SwitchSwitch

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0.5

1

1.5

2

2.5

IL (d

B)

Insertion Losses (ON)


(b)

Figure F.21: Isolator + switch: (a) Insertion losses when switch is OFFand (b) insertion losses when the switch is ON.

Figure F.20a shows that input return losses in the off-state have improved by 15 dB withrespect to the return losses provided by the switch alone. This is compliant with the 20 dBreverse attenuation and 14 dB input reflection coefficient provided by the isolator. In both, off-and on-states, reflections coming from the switch or the LNA can be neglected in comparisonwith those directly connected to the isolator reflection coefficient.

At the same time, insertion losses in the on-state (Figure F.21b) have not been severelyworsened. The switch alone has 1 dB insertion loss and the isolator, 1.3 dB. Both devicestogether obtain 2.2 - 2.5 dB insertion losses in the band of interest, as expected.

As a whole, the ensemble works as an absorptive switch, which is a device that is not currentlycommercialized at W-band. Table F.7 summarizes the obtained measurement results.


Parameter Measured

Insertion losses (S21 ON) 2 dB

Isolation (S21 OFF) 35.3 dB


Table F.7: Measurement results of the switch and the isolator at 94GHz.

F.7 RPG W-LNA75110 Low Noise Amplifier

The LNA is the first active device of the receiver chain. It should be located as close as possibleto the antenna to reduce the noise figure of the chain. Figure F.22 shows the measurementset-up that has been used for the characterization of this LNA.

Figure F.22: Measurement set-up for the RPG W-LNA75110 low noiseamplifier.

Measured S-parameters are presented in Figure F.23 and gain and return losses can beobserved in detail in Figure F.24.

85 90 95 100 105

Frequency (GHz)

-50

-40

-30

-20

-10

0

10

20

30

(dB

)

S-Parameters

S11S12S21S22

(a)

85 90 95 100 105

Frequency (GHz)

-150

-100

-50

0

50

100

150

(º)

S-Parameters

S11S21

(b)

Figure F.23: RPG W-LNA75110 low noise amplifier: (a) S-Parametersamplitude and (b) S-Parameters phase.

124 F.7. RPG W-LNA75110 LOW NOISE AMPLIFIER

90 91 92 93 94 95 96 97 98

Frequency (GHz)

20

25

30

35

G (d

B)

Gain

MeasuredDatasheet

(a)

90 91 92 93 94 95 96 97 98

Frequency (GHz)

0

5

10

15

RL

(dB

)

Return Losses

Input PortOutput PortInput DatasheetOutput Datasheet

(b)

Figure F.24: RPG W-LNA75110 low noise amplifier: (a) Gain(20 log|S21|) and (b) return losses.

This device features a gain (Figure F.24a) of 26 - 31 dB in the band of interest, which isslightly above the manufacturer curve. Input return losses (Figure F.24b) are 8.4 - 11.2 dB. Allthis is compliant with the specification. It is also important to note that power rating of thisdevice is �20 dBm, so a complete protecting network will be needed at its input to isolate itfrom the transmitter power.



Gain 28 dB 26.5 dB

Input return losses 9.2 dB 9.3 dB

Table F.8: RPG W-LNA75110 low noise amplifier measurement resultsat 94GHz in comparison with those provided by the manufacturer.

Appendix G

Millimeter Wave ComponentsDatasheets

The datasheets collected in this appendix have been organized as follows:

1. RF components:

• Passive devices:

– RPG WFI-110 isolator.

– RPG WPD-110 hybrid power divider.

– ELVA CR-1094 circulator.

– Quinstar QAL-W00000 variable attenuator.

– ELVA SPST-10 switch.

• Active devices:

– RPG W-LNA75110 low noise amplifier.

– Quinstar QMB-9999WS balanced mixer.

– RPG AFM6-110 ×6 frequency multiplier.

2. Optical structures:

• Edmund Optics aluminium off-axis mirrors.

– 84-585 45◦ off-axis paraboloid mirror.



• W-band Teflon lenses.

3. Antennas:

• Millitech SGH-08 conical horn in WR-08.

• RPG FH-PP-100 Potter horn in WR-10.

125

APPENDIX G. MILLIMETER WAVE COMPONENTS DATASHEETS 127

128

26.5-220 GHz operating frequency Narrow band and full band types Low insertion losses

High isolation Compact size

Laboratory measurement and test equipment Junction of some parts of sub-systems Matching of several waveguide components Base of multi-junction devices

(injection-locked amplifier)

26.5-220GHz Circulators and Isolators are ferrite waveguide components. There are two kinds of

the products operating within narrow frequency band (1-4 GHz) and full waveguide band. Junction circulators /isolators are narrow band. Wideband devices base on Faraday rotation effect. Used in many waveguide schemes for junction their parts and for matching different components and protecting against reflected mm-wave power.

Standard line of ELVA-1’s circulators CR-XX/CF/BW series provide low insertion losses and high isolation for all three ports. They have operation frequency band up to 4 GHz. Better performances of the circulators can be provided within narrow frequency band.

The IS-XX/CF/BW isolators realized by terminating of one port of the junction circulators. Ideally suit for suppression of reflected power coming from any waveguide devices with high VSWR.

Full band isolators IF-XX have good performances within full waveguide range. Mainly used in wideband sources or receivers for suppression mm-wave power propagating in one fixed direction. Have small insertion losses.

Narrow band circulators CR-XX/CF/BW series:

CR-XX/CF/BW Central frequency: Fixed from 26.5- 110 GHz Fixed from 110- 170 GHz Bandwidth: 2.5 GHz 0.5 GHz 2.5 GHz 0.5 GHz Insertion losses: 1 to 2, 2 to 3, 3 to 1 0.7 dB 0.5 dB 1 dB 0.7 dB

Isolation: 2 to 1, 3 to 2, 1 to 3 23 dB (min) 30 dB 20 dB (min) 25 dB

VSWR 1.2 (typ) 1.3 (typ)

1 2

3

130

Low insertion losses High isolation Low cost

Fast switching time More then 10% bandwidth operation Easy to use

Radars Fast protection system AM of microwave signals. Lock-in detection systems

ELVA-1 series fast SPST switches is built on slim film PIN

diodes. Built-in driver provides switching time 4-6 ns and unique technology allows to get more then 10% operation with small insertion losses and isolation more then 30 dB.

Supply +/-5V DC Control signal TTL Control Input impedance 50 Ohm *The models with 60 dB Isolation are available upon request **Guaranteed for Rise Time 0-90% RF and Fall Time 100-10% RF. Typical data for different models are presented below.

Specify Model Number SPST-XX/AA/BB - XX- waveguide band (WR-Number) - AA – Center operation frequency (fo), GHz - BB – Operation bandwidth (fo+/-BB), GHz

Model SPST-42 SPST-28 SPST-22 SPST-19 SPST-15 SPST-12 SPST-10 SPST-08 SPST-06 Frequency Band K Ka Q U V E W F D Range, GHz 18-26.5 26-40 33-50 40-60 50-75 60-90 75-110 90-140 110-150 Insertion Loss, dB (typ) 0,7 0,7 0,8 0,8 0,8 1,0 1,0 1,5 1,5 Isolation, dB (min) 30* 30* 30* 30* 30* 30* 30* 30* 30* Peak Power, W(max) 1,0 1,0 1,0 1,0 1,0 1,0 1,0 1,0 0,8 Switching Time, ns** 4-6 4-6 4-6 4-6 4-6 4-6 4-6 4-6 4-6

SPST-10/94

-40

-35-30

-25-20

-15

-10-5

0

90 90,5 91 91,5 92 92,5 93 93,5 94 94,5 95 95,5 96 96,5 97 97,5 98 98,5 99 99,5 100

Frequency, GHz

Loss

, dB

'0', Isolation '1', Loss Switch SPST-06/140

-50-45-40-35-30-25-20-15-10-50

110

112

114

116

118

120

122

124

126

128

130

132

134

136

138

140

142

144

146

148

150

Frequency, GHz

Loss

, dB

Isolation Losses

136

W-Band Dielectric Lenses

Parameter Value

Material Teflon (εr = 2.1, tan(δ) = 0.006)

Focal length 62.7mm

Input radius ∞Output radius 26.961mm

Lens thickness 21.41mm

Losses 19.6%

Back-vertex focal length 48mm

Front-vertex focal length 62.7mm


SERIES SGH Millimeter-Wave Technology & Solutions

ELECTRICAL SPECIFICATIONS*

Pyramidal HornsModel Number SGH-42 SGH-28 SGH-22 SGH-19 SGH-15 SGH-12 SGH-10 SGH-08 SGH-06 SGH-05 SGH-04 SGH-03Frequency band and range (GHz)

K18-26.5

Ka26.5-40

Q33-50

U40-60

V50-75

E60-90

W75-110

F90-140

D110-170

G140-220

-170-260

-220-325

Gain (dB) 24 24 24 24 24 24 24 24 24 24 24 24VSWR 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.25:1 1.25:1 1.25:1 1.25:1 1.25:1

Conical HornsGain (dB) 21 21 21 21 21 21 21 21 21 21 21 21VSWR 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.2:1 1.25:1 1.25:1 1.25:1 1.25:1 1.25:1*All specifications listed are typical values.

TYPICAL PERFORMANCE

OUTLINE DRAWINGS*

*The outlines shown may not reflect the latest information. Please contact Millitech for current outline drawings.

IS000025 REV07 ECO #1606-29-03www.millitech.com

138

SERIES SGH Millimeter-Wave Technology & Solutions

MECHANICAL SPECIFICATIONS

Circular Waveguide Output – Conical HornsDiameter (in) Frequency Range A (in/mm) B (in/mm) Flange MIL.F-3922

0.455 17.5-20.5 2.300/58.420 5.000/127.000 /54-001*0.396 20.0-24.5 2.660-67.564 4.976/126.390 /54-001*0.328 24.0-26.5 2.350/59.690 4.384/111.354 /54-001*0.315 26.5-33.0 1.800/45.720 3.300/83.820 /54-003*0.250 33.0-38.5 1.660/42.164 3.097/78.664 /54-003*0.219 38.5-40.0 1.512/38.405 2.821/71.653 /54-003*0.250 33.0-38.5 1.440/36.576 2.700/68.580 /67B-0060.219 38.5-43.0 1.456/39.982 2717/69.012 /67B-0060.188 43.0-50.0 1.276/32.410 2.381/60.477 /67B-0060.210 40.0-43.0 1.180/29.972 2.250/57.150 /67B-0070.188 43.0-50.0 1.276/32.410 2.381/60.477 /67B-0070.165 50.0-60.0 1.079/27.407 2.013/51.130 /67B-0070.165 50.0-58.0 0.950/24.130 1.900/48.260 /67B-0080.141 58.0-68.0 0.942/23.927 1.757/44.628 /67B-0080.125 68.0-75.0 0.830/21.082 1.548/39.319 /67B-0080.136 60.0-66.0 0.784/19.914 1.600/40.640 /67B-0090.125 66.0-88.0 0.770/19.558 1.438/36.525 /67B-0090.094 88.0-90.0 0.667/16.942 1.244/31.598 /67B-0090.112 75.0-88.0 0.644/16.358 1.300/33.020 /67B-0100.094 88.0-110.0 0.599/15.215 1.118/28.397 /67B-0100.089 90.0-115.0 0.514/13.056 1.100/27.940 /67B-M080.075 115.0-140.0 0.465/11.811 0.868/22.047 /67B-M080.073 110.0-140.0 0.418/10.617 0.900/22.860 /67B-M060.059 140.0-160.0 0.396/10.058 0.738/18.745 /67B-M060.058 140.0-220.0 0.328/8.331 0.750/19.050 /67B-M050.049 170.0-260.0 0.276/7.010 0.650/16.510 /67B-M040.039 220.0-325.0 0.218/5.537 0.550/13.970 ---

Rectangular Waveguide Output – Pyramidal HornsModel Number A (in/mm) B (in/mm) C (in/mm) Flange MIL.F-3922

SGH-42 4.068/103.327 3.093/78.562 7.480/189.992 /54-001*SGH-28 2.712/68.885 2.062/52.375 5.087/129.210 /54-003*SGH-22 2.170/55.118 1.650/41.910 4.070/103.378 /67B-006SGH-19 1.821/46.253 1.385/35.179 3.480/88.392 /67B-007SGH-15 1.434/36.424 1.090/27.686 2.775/70.485 /67B-008SGH-12 1.182/30.023 0.898/22.809 2.348/59.639 /67B-009SGH-10 0.969/24.613 0.736/18.694 1.938/49.225 /67B-010SGH-08 0.775/19.685 0.589/14.961 1.561/39.649 /67B-M08SGH-06 0.630/16.002 0.479/12.167 1.265/32..131 /67B-M06SGH-05 0.494/12.548 0.376/9.550 1.036/26.314 /67B-M05SGH-04 0.417/10.592 0.317/8.052 0.855/21.717 /67B-M04SGH-03 0.329/8.357 0.250/6.350 0.707/17.958 ---

* With #4-40 threaded holes.

IS000025 REV07 ECO #1606-29-03www.millitech.com

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