A Two-Pole Lumped-Element Programmable Filter With MEMS Pseudodigital Capacitor Banks

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 56, NO. 3, MARCH 2008 729

A Two-Pole Lumped-Element Programmable FilterWith MEMS Pseudodigital Capacitor Banks

Cristiano Palego, Associate Member, IEEE, Arnaud Pothier,Aurelian Crunteanu, Matthieu Chatras, Pierre Blondy, Member, IEEE,

Corinne Champeaux, Pascal Tristant, and Alain Catherinot

Abstract—This paper presents a novel two-pole reconfigurablebandpass filter on alumina substrate for applications in - and

-bands. An analytical approach was followed for synthesis of mul-tiple filtering characteristics. A microstrip network on alumina wasthen optimized to implement a set of switched filtering functions.Finally a reconfigurable filter was fabricated and tested in order tovalidate the proposed approach. This filter exploits five 3-bit capac-itor banks controlled with microelectromechanical systems ohmicswitches to achieve 12 states with 37.5% tuning range between1.51–2.26 GHz. Several tuning mechanisms are demonstrated in-cluding frequency, bandwidth tuning, and frequency+ bandwidthtuning. Good agreement with theoretical results has been obtained.

Index Terms—Filters, RF microelectromechanical systems(MEMS), tunable filters.

I. INTRODUCTION

THE LATEST generation of wireless and radar communi-cation systems shows increasing interest in reconfigurable

circuits with high-sensitivity and low-complexity receivers.RF microelectromechanical systems (MEMS) componentsoffer ideal tuning solutions for front-ends due to linear high-low-intermodulation behavior combined with virtually no dcpower consumption. MEMS tunable filters, reconfigurable bothin bandwidth and in central frequency, have been successfullydeveloped using capacitive or ohmic switches [1]–[4]. Recentresearch have demonstrated pseudodigital frequency tuning [3],[4], which increases immunity to noise, reduces temperaturesensitivity, and achieves better performance reproducibility.

Most results available today are on MEMS bandpass filterson silicon, quartz, or glass substrates. However, implementationof MEMS technology on alumina substrates would allow ex-tremely compact designs, low loss, and simple packaging [5].Furthermore, this would increase the already remarkable inte-gration potential of MEMS components to the many receiversubsystems that are fabricated on ceramic substrates.

In this paper, we present a two-pole reconfigurable filterusing MEMS series ohmic switches [6]. They are used to

Manuscript received March 17, 2007; revised October 2, 2007.C. Palego was with the XLIM Research Institute, National de la Recherche

Scientifique (CNRS), University of Limoges, 87060 Limoges Cedex, France.He is now with the ECE-CSTL Department, Lehigh University, Bethlehem, PA18015 USA.

A. Pothier, A. Crunteanu, M. Chatras, P. Blondy, C. Champeaux, P. Tris-tant, and A. Catherinot are with the XLIM Research Institute, National de laRecherche Scientifique (CNRS), University of Limoges, 87060 Limoges Cedex,France (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2008.916873

Fig. 1. Circuit model for the electric coupling of two synchronous resonators.

obtain discrete capacitance values from five lumped capacitorbanks. In addition, two surface-mounted inductors are com-bined with two capacitor banks, providing a pair of tunablelumped high- resonators. The lumped-element implementa-tion makes the final design very compact and prevents problemswith spurious harmonics that typically affect distributed solu-tions. Finally, this architecture provides a large number of statesfor the reconfigurable filter. Such a tunable network operatesas a programmable filter achieving multiple filtering functionsusing different tuning mechanisms.

II. FILTER DESIGN

A. Design Principle

Coupled resonator circuits are extensively being used inmicrowave/RF filter design, especially for narrowband filters,and allow effective integration of MEMS components to planarstructures. Indeed, MEMS tunable capacitances can easily beused to control the electric coupling of planar resonators, aswell as their resonant frequency. General synthesis techniqueshave been developed based on coupling coefficient of inter-cou-pled resonators and external quality factor of the input/output(I/O) resonators [7]–[9]. Electric coupling is achieved usingmutual capacitances between resonators and is well suited forpractical implementation with MEMS tunable capacitances.This provides effective and simple control on coupling co-efficients, while tuning magnetic couplings would be verycomplex, bulky, and expensive to implement.

In the case of two electrically coupled resonators, an equiv-alent lumped model is provided in Fig. 1 [8], where identicalresonators are used in a synchronously tuned filter. andare, respectively, the inductance, capacitance, and source con-ductance, while is the mutual coupling capacitance. Theelectric coupling between two resonant loops can be represented

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730 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 56, NO. 3, MARCH 2008

Fig. 2. Lumped-element final model for the tunable filter with practical I/Oimpedance transformer.

by an admittance inverter [8]. This reduces the cou-pled resonator structure to the standard bandpass filter sectionwith parallel resonators and J-inverters. This also allows a sys-tematic synthesis based on the insertion-loss method [9].

The coupling coefficient for the two coupled-resonator net-work in Fig. 1 is commonly defined as

(1)

The two resonators are connected to the I/O. Their externalquality factor is

(2)

where is the resonance frequency.The conductance -parameter corresponds to the network

admittance and it is one of the tuning parameters that insuresfeasibility of any second-order filtering function. Furthermore,setting allows to keep the inductance to a constant value re-gardless of the filter transfer function. Doing so, no multiple ortunable inductors are needed and a multifunction filter can beimplemented with a single and simple architecture. The valuecan also be chosen relatively large ( nH) in order to obtaincompact LC resonators with feasible capacitors ( pF).

In practice, tunable impedance inverters are integrated tocouple and match the filter to the external 50- feeding lines.These elements are also implemented using MEMS variablecapacitances, as shown in Fig. 2.

The ratio of the input/resonator capacitances for theimpedance scaled filter in Fig. 2 is derived by network analysisfrom the original circuit (Fig. 1) around

(3)

By algebraic manipulations, the filter key parameters andcan be directly expressed as a function of the MEMS tunable

and capacitances

(4)

MEMS-based capacitor banks are used to achieve discretetuning of and components. Different sets of ca-pacitive values can be selected to implement different filtering

functions. Hence, by properly choosing the achievable capaci-tive values, the filter can be programmed to perform a variety offrequency responses with a great degree of flexibility in centralfrequency and bandwidth.

B. Implementation of Multiple Filtering Functions

An implementation procedure is done following the standardinsertion-loss method [9] and consists of determining the un-known - - and -parameters from the following three re-lations:

(5)

where and are the filter specified center frequencyand the fractional bandwidth, respectively. and are theChebyshev coefficients for a second-order equiripple response.The inductance nH is chosen among commerciallyavailable components, and is a constant parameter for every fil-tering function. A common return-loss level (15 dB) at is alsoset, and the 3-dB fractional bandwidth is used, which is conve-nient, in practice, to measure. This only implies slight modifi-cations in (5) so that several filtering functions can be rapidlyimplemented by simple normalization/de-normalization opera-tions. Finally, the impedance scaled filter - and -parame-ters are easily derived from (3).

For instance, if two identical LC resonators are coupled atGHz with % and the desired return loss is

15 dB, then from (1)–(5),

pF

pF

pF

pF.

These parameters are those of the sixth function of Table I,while the resulting frequency response corresponds to thethicker curves in Figs. 3–5.

Fig. 3 shows the influence of and capacitanceson the external quality factor and coupling coefficient.and curves are parameterized by increasing values of

and couples, corresponding to identicalChebyshev filter prototypes with different normalization. Thehighlighted points correspond to particular andtriplets that are being used in this study. The present MEMStunable capacitor banks allow to set the filter parameters to anyof these points, resulting in a combined discrete variation of

- and -parameters, corresponding to the same Chebyshevprototype. This corresponds to discrete tuning of the normal-ization of this prototype regarding bandwidth or/and centralfrequency parameters. Changing normalization corresponds tothree distinct tuning mechanisms that can be set up to obtainmultiple responses.

A first set of nine filtering functions have been designed touniformly cover the 1.5–2.3-GHz frequency band with constant

PALEGO et al.: TWO-POLE LUMPED-ELEMENT PROGRAMMABLE FILTER WITH MEMS PSEUDODIGITAL CAPACITOR BANKS 731

TABLE IMEASURED FILTERING FUNCTIONS

Fig. 3. Q andK as a function of C - and C -parameters.

Fig. 4. Multiple response simulation: frequency tuning.

absolute bandwidth GHz. Thus, varies as ,while increases with , and the resulting responses ex-tend over the frequency axis with constant insertion loss and

Fig. 5. Multiple response simulation: bandwidth tuning.

return loss. This implements frequency tuning of the frequencyresponse over the chosen band, as can be seen in Fig. 4, whilethe corresponding parameters are reported in the first nine rowsof Table I.

Functions 10 and 11 in Table I allow to perform bandwidthtuning of the response around the reference frequency. thenvaries as is kept constant, and bandwidth is directly con-trolled by the coupling capacitance value while the otherfiltering properties remain unchanged. Bandwidth tuning char-acteristics are shown in Fig. 5.

Finally, frequency and bandwidth tuning can be combined sothat and are increasing linearly together. This is referredto as mixed tuning; the corresponding curves are reported in thelast three rows of Table I and appear in Fig. 6.

C. Lumped Tunable Resonators Microstrip Design

The filter is implemented on a 256- m-thick aluminasubstrate using two high- integrated inductors (Coilcraft,

– nH, – – GHz,GHz) and lumped pseudodigital capacitor banks

[4].Each fixed value capacitor bit can be connected in parallel to

others bank using MEMS switches, resulting in discrete capaci-tance variation. All variable capacitors consist of the series com-bination of MEMS ohmic switches and metal–air–metal capac-itors for capacitances, or printed microstrip capacitorsfor capacitances. The combination between different typesof bit and MEMS ohmic switches is illustrated in Fig. 7 showing


Fig. 6. Multiple response simulation: mixed tuning.

Fig. 7. Different capacitive banks including either metal–air–metal or printedmicrostrip capacitors.

how switches are used in pairs or triplets to reduce the resultingohmic contact loss.

The 3-bit capacitor banks are controlled by three independentpairs or triplets of switches and provide 2 possible capacitivestates. This is achieved by changing the surface of the fixed ca-pacitors. For example, in the and capacitor banks shownin Fig. 8, different weights are achieved by changing the surfacearea of the lower electrode of metal–air–metal series capac-itors. Similarly, in the banks, the parallel-plate printed mi-crostrip capacitors have different surface areas. The weights arechosen so that their combination results in a piecewise linear ca-pacitance variation, as shown in Fig. 8.

The achievable capacitance values and their quality factor areextracted by Agilent Momentum electromagnetic (EM) simula-tions that allow computing the discrete capacitor value for eachbank. MEMS switch ohmic losses are included in simulations,assuming a global resistance series resistanceresulting from 1- contact resistance per single contact in theactuated state. The corresponding factor depends on bit com-binations and the simulated minimum values are reportedin Fig. 8 for each bank.

In practice, MEMS capacitance sets are not fully indepen-dent because of the presence of transmission lines and biasinglines. Therefore, careful EM design is needed to properly ar-range the bit combinations and to deal with parasitic inductiveeffects from inter-capacitor connection with nonzero electricallength.

Fig. 8. Different bit configuration in the C ;C ; and C capacitor banksdiscrete achievable values.

Fig. 9. MEMS switch control on a metal–air–metal capacitor discrete achiev-able values.

The surface-mounted inductor effect is taken into account byhybrid circuit simulations. The geometry of the filter is opti-mized for the Coilcraft inductors size and shape. Indeed, as canbe seen from Fig. 7, the capacitor banks are arranged around thesurface-mounted components in order to obtain very compacttunable resonators. Furthermore, the inductors bonding pads areused as shunt capacitors through the alumina substrate, and areincluded in .

Finally, Fig. 9 reports details of the filter biasing network,while the whole set of optimized sates is visible in Fig. 10. Eachbiasing line ends in a contact pad including an integrated resistorto reduce parasitic coupling between the MEMS switches andthe biasing system. Integrated resistors are also being used toapply dc potential between different bits and minimize RF signalleakage in the biasing network. It can be seen in Figs. 9 and11 that the number of components included in such a biasingnetwork is considerable. This results in high sensitivity of thefilter electric performance to the value of biasing resistance thatis actually achieved in the fabrication process, as shown at theconclusion of this paper.


Fig. 10. EM simulated S-parameters for the reconfigurable filter. (a) Fre-quency tuning. (b) Bandwidth and mixed tuning.

Fig. 11. Microphotograph of the fabricated filter.

III. FABRICATION AND MEASUREMENT

A. Fabrication

The circuits have been fabricated on a 256- m mirror pol-ished alumina substrate. The backside metallization is madewith a 3- m-thick electroplated gold. 1-k /square SiCr resis-tors are thermally evaporated on the front side and patterned

Fig. 12. Measured insertion loss and return loss for the frequency tuned states.

using a liftoff technique.This is followed by a 200/1000-ÅCr/Au layer evaporation to form the filter biasing network. Inthe next step, a 3500-Å-thick pulsed laser deposition (PLD)alumina insulating layer is deposited and patterned to pro-tect the biasing network. A 200/1500-Å Cr/Au layer is thenevaporated and electroplated up to 1 m in order to form thefilter first metallization layer and the switch contact fingers.Next, a 1.8- m-thick polymer is used as a sacrificial layerfor the MEMS cantilevers and patterned in two steps to etchcontact dimples. The top electrode for MEMS switches and allcapacitors is created by first evaporating a seed layer of Ti/Au(100/1500 Å) and then electroplated up to 3 m to obtain stiffcantilevers structures and low-loss metallization.

Once the second metal layer is patterned, each filter is indi-vidually diced using a precision wire saw in order to separatethe circuits to be measured. The sacrificial layer is then removedto release MEMS devices. Finally the integrated inductors arealigned and bonded to their pads using silver epoxy glue. Thesame epoxy is used to put the circuit on a metallic support formeasurement and make via connections to ground for inductorsand I/O coplanar waveguide (CPW) to microstrip transitions.

B. Measurements

Measured insertion and return loss are shown in Fig. 12–14.Measurements are performed on-wafer by a short-open-load-thru calibration technique, which includes loss due to I/O CPWto microstrip RF transitions.

The measured insertion-loss ranges from 2.9 dB, for thebroadband state at 2.03 GHz, to 5.9 dB for the lowest frequencystate at 1.51 GHz. The return loss are always better than 14 dB.It can be seen that the measured responses are in good agree-ment with the EM and circuit simulations ones.

As can be seen, the frequency distribution is not as perfectas designed, but center frequency deviation from simulations tomeasurements never exceed 2.4%, as is observable from Table I.


Fig. 13. Measured insertion and return loss for bandwidth tuning.

Fig. 14. Measured insertion and return loss for mixed frequency/bandwidthtuning.

The reference state is located at 1.98 GHz with 10.5% , whichis very close to simulated results.

Regarding frequency tuning, (Fig. 12), the measured isclose to simulated ones, although a bit broader for higher fre-quency states.

This corresponds to a moderate drift in the extractedand values. The highest frequency state turns out to be at2.26 GHz and sets the overall tunability to 37.5%, close to thetheoretical value.

The bandwidth tuning performance is close to EM simula-tions and can be observed in Fig. 13 where the reference stateis compared to the largest 0.4 GHz state at 2.03 GHz. Mixedtuning measured responses are visible in Fig. 14 and also showgood agreement with simulated parameters.

Fig. 15. Simulated loss performance for different biasing resistance values at1.51 GHz.

Fig. 16. Resonator Q-factor evolution as a function of the bias line resistivityr .

Section III-C provides a simple interpretation of the measuredinsertion-loss performance by modeling the biasing network in-fluence.

C. Insertion-Loss Performance Improvement

This filter exhibits higher measured insertion loss than ex-pected and it is a parameter that can be improved. It has beenshown how insertion loss can be improved by increasing theresistivity of the bias line integrated resistors [3]. The effec-tive resistivity value achieved during fabrication is a difficultparameter to control and has not been measured. However, hy-brid EM/circuit simulations have been performed on the filter,including different biasing resistance value, and accurately re-produce the measured loss performance for k /square.Fig. 15 compares measured and simulated scattering parame-ters for the state at 1.51 GHz (worst state performance) and fordifferent values. It can be noticed that the insertion-loss per-formance reaches 2.1 dB for 100-k /square resistivity.

The extracted resonator factor resulting from the interac-tion with the biasing network is for k /squarein measurements. Fig. 16 shows the -factor evolution fordifferent values. quickly improves as is increasedto 10 k /square, and then slowly approaches forhigh-resistivity levels ( k /square).

IV. CONCLUSION

A new type of reconfigurable filter on alumina has beendesigned, fabricated, and tested. This filter achieves recon-figurability between multiple filtering functions combining


different tuning mechanisms. Measurement results have val-idated the principle of reconfiguration and demonstrated12-state operations with an overall tuning range of 37.5% inthe 1.51–2.26-GHz band. The desired filtering functions can beachieved in good agreement with simulations and the deviceadditionally exhibits very compact size, making it suitable forminiaturized applications.

These encouraging results will allow further developmentof reconfigurable devices on an alumina substrate that will bebased on the same principle with improved frequency resolu-tion and tunability. There would also be considerable interestin generalizing the presented technique to -coupled resonatornetworks. This would provide control over the out-of-bandrejection of multiple filtering characteristics and would allowimplementation of programmable filters, where central fre-quency, bandwidth, and selectivity can be simultaneously set.

REFERENCES

[1] E. Fourn et al., “MEMS switchable interdigital coplanar filter,” IEEETrans. Microw. Theory Tech., vol. 51, no. 1, pp. 320–323, Jan. 2003.

[2] R. M. Young et al., “Low-loss bandpass RF filter using MEMS ca-pacitance switches to achieve a one-octave tuning range and indepen-dently variable bandwidth,” IEEE MTT-S Int. Microw. Symp. Dig., pp.1781–1784, 2003.

[3] K. Entesari and G. M. Rebeiz, “A 12–18 GHz three 8-pole RF MEMStunable filter,” IEEE Trans. Microw. Theory Tech., vol. 53, no. , pp.2566–2571, Aug. 2005.

[4] A. Pothier et al., “Low loss two bit tunable bandpass filter using MEMSDC contact switches,” IEEE Trans. Microw. Theory Tech., vol. 53, no.1, pp. 126–129, Jan. 2005.

[5] J. Mckillop, T. Fowler, D. Goins, and R. Nelson, “Design, performanceand qualification of a commercially available MEMS switch,” in 36thEur. Microw. Week Conf. Dig., Manchester, U.K., Sep. 2006, pp.1399–1401.

[6] A. Pothier et al., “MEMS DC contact micro relays on ceramic substratefor space communication switching networks,” in 35th Eur. MicrowaveWeek Conf. Dig., Paris, France, Oct. 2005, pp. 629–632.

[7] R. J. Cameron, “General coupling matrix synthesis methods for Cheby-shev filtering functions,” IEEE Trans. Microw. Theory Tech., vol. 47,no. 4, pp. 433–442, Apr. 1999.

[8] J.-S. Hong and M. J. Lancaster, Microstrip Filters for RF/MicrowaveApplications. New York: Wiley, 2001.

[9] A. E. Atia and A. E. Williams, “Narrow-bandpass waveguide filters,”IEEE Trans. Microw. Theory Tech., vol. MTT-20, no. 4, pp. 258–265,Apr. 1972.

[10] G. L. Matthaei, L. Young, and E. M. T. Jones, Microstrip Filters,Impedance-Matching Networks and Coupling Structures. Dedham,MA: Artech House, 1980.

Cristiano Palego (S’03–A’06) received the Ph.D. degree from the Universityof Limoges, Limoges, France, in 2006.

In 2007, he joined the ECE-CSTL Department, Lehigh University, Beth-lehem, PA, where he is a currently a Post-Doctoral Fellow. His main researchinterests are in MEMS circuits design and modeling.

Arnaud Pothier received the Ph.D. degree from theUniversity of Limoges, Limoges France, in 2003.

In 2005, he became a Full-Time Researcher withthe Centre National de la Recherche Scientifique(CNRS), XLIM Research Institute, University ofLimoges. His main research interests are MEMS cir-cuits design and modeling, and biological interactionwith EM waves at microscale/nanoscale.

Aurelian Crunteanu received the Ph.D. degree fromthe University of Bucharest, Bucharest, Romania, in2002.

In 2004, he became a Centre National de laRecherche Scientifique (CNRS) Researcher with theXLIM Research Institute, University of Limoges.His current research interests include RF-MEMS,MOEMS, and applications of new materials to RFsystems/subsystems.

Matthieu Chatras received the Ph.D. degree from the University of Limoges,Limoges, France, in 2003.

In 2004, he became an Assistant Professor with the University of Limoges.His current research interests include RF MEMS, micromachining, and appli-cations of piezoelectric materials at microwaves.

Pierre Blondy (M’00) received the Ph.D. and Ha-bilitation degrees from the University of Limoges,Limoges, France, in 1998 and 2003, respectively.

In 1998, he joined the Centre National de laRecherche Scientifique (CNRS), XLIM ResearchInstitute, University of Limoges. In 2006, he becamea Professor with the University of Limoges, where heconducts research on microwave filters, RF MEMS,and EM-based design of microwave circuits.

Dr. Blondy is a member of the IEEE MicrowaveTheory and Techniques Society (IEEE MTT-S) and

the European Microwave Association (EuMa) Society. He serves on the Tech-nical Program Committee of the IEEE MTT-S International Microwave Sym-posium (IMS) since 2003.

Corinne Champeaux received the Ph.D. degreein electrical engineering from the University ofLimoges, Limoges, France, in 1992.

Since 1992, she has been an Assistant Professorwith the Faculty of Science, University of Limoges.She currently conducts research with the Sciencesdes Procédés Céramiques et de Traitements de Sur-face (SPCTS) Laboratory, Unité Mixte de Recherche(UMR) 6638, Centre National de la RechercheScientifique (CNRS), University of Limoges. Hermain research interests are laser–matter interactions

and pulsed-laser thin-films deposition techniques. She is involved with thedevelopment and fabrication of MEMS components through the elaboration ofnew materials and fabrication processes.

Pascal Tristant received the Ph.D. degree from the University of Limoges,Limoges, France, in 1991.

In 1991, he became an Assistant Professor with the University of Limoges.His current research interests include PECVD optimization for ceramic deposi-tion techniques and optimization.

Alain Catherinot received the Doctorate degree inphysical sciences from the University of Orléans, Or-léans, France, in 1980.

He is currently a Professor with the Processand Material Sciences Department, University ofLimoges, Limoges, France, where he conductsresearch on PLD techniques with the Sciences desProcédés Céramiques et de Traitements de Surface(SPCTS) Laboratory, Unité Mixte de Recherche(UMR), Centre National de la Recherche Scien-tifique (CNRS), University of Limoges. His research

interests include plasma and laser materials interactions, PLD of thin films andnanostructured materials, and their characterizations. He is also involved inMEMS fabrication using innovative deposition methods and materials.

A Two-Pole Lumped-Element Programmable Filter With MEMS Pseudodigital Capacitor Banks

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