Electro-optic modulator with exceptional power-size performance enabled by transparent conducting...

Electro-optic modulator with exceptional power-

size performance enabled by transparent

conducting electrodes

Fei Yi1., Fang Ou

1., Boyang Liu

1., Yingyan Huang

1., Seng-Tiong Ho

1*, Yiliang Wang

2.,

Jun Liu2., Tobin J. Marks

2., Su Huang

3., Jingdong Luo

3., Alex K.-Y. Jen

3., Raluca Dinu

4.

and Dan Jin4.

1Department of Electrical Engineering and Computer Science, Northwestern University, 2145 Sheridan Rd,

Evanston, IL, 60208, USA 2Department of Chemistry, Materials Research Center, Northwestern University, 2145 Sheridan Rd, Evanston IL,

60208, USA 3Department of Materials Science and Engineering, Box 352120, University of Washington, Seattle, WA, 98195, USA

4GigOptix Inc, 2400 Geng Road, Suite 100, Palo Alto, CA. 94303, USA

*[email protected]

Abstract: An EO phase modulator having transparent conducting oxide

electrodes and an inverted rib waveguide structure is demonstrated. This

new modulator geometry employs an EO polymer having an in-device r33 =

60pm/V. The measured half-wave voltage Vπ of these devices ranges from

5.3V to 11.2V for 3.8 and 1.5 mm long devices, respectively. The lowest

VπL figure-of-merit corresponds to 0.6V-cm (7.2mW-cm2 of power length

product) in a dual-drive configuration. The trade-off between Vπ, insertion

loss and modulation bandwidth is systematically analyzed. An optimized

high-speed structure is proposed, with numerical simulation showing that

this new structure and an in-device r33 = 150pm/V, can achieve Vπ = 0.5V

in a 5mm long active length with dual drive operation. The insertion loss is

targeted at 6dB, and a 3dB optical modulation bandwidth can reach >

40GHz.

©2010 Optical Society of America

OCIS codes: (250.4110) Modulators; (250.2080) Polymer active devices.

References and links

1. W. S. C. Chang, RF Photonics Technology in Optical Fiber Links, Cambridge University Press (2002).

2. M. Haurylau, G. Chen, H. Chen, J. Zhang, N. A. Nelson, D. H. Albonesi, E. G. Friedman, and P. M. Fauchet,

“On-chip Optical Interconnect Roadmap: Challenges and Critical Directions,” IEEE J. Sel. Top. Quantum

Electron. 12(6), 1699–1705 (2006).

3. B. Kuhlow, “Modulators,” in Laser Fundamentals Part 2, Springer Berlin Heidelberg (2006).

4. Y.-J. Cheng, J. Luo, S. Huang, X. Zhou, Z. Shi, T.-D. Kim, D. H. Bale, S. Takahashi, A. Yick, B. M. Polishak,

S.-H. Jang, L. R. Dalton, P. J. Reid, W. H. Steier, and A. K.-Y. Jen, “Donor-Acceptor Thiolated Polyenic

Chromophores Exhibiting Large Optical Nonlinearity and Excellent Photostability,” Chem. Mater. 20(15),

5047–5054 (2008).

5. P. A. Sullivan, H. Rommel, Y. Liao, B. C. Olbricht, A. J. P. Akelaitis, K. A. Firestone, J.-W. Kang, J. Luo, J. A.

Davies, D.-H. Choi, B. E. Eichinger, P. J. Reid, A. Chen, A. K.-Y. Jen, B. H. Robinson, and L. R. Dalton,

“Theory-guided design and synthesis of multichromophore dendrimers: an analysis of the electro-optic effect,” J.

Am. Chem. Soc. 129(24), 7523–7530 (2007).

6. T.-D. Kim, J.-W. Kang, J. Luo, S.-H. Jang, J.-W. Ka, N. Tucker, J. B. Benedict, L. R. Dalton, T. Gray, R. M.

Overney, D. H. Park, W. N. Herman, and A. K.-Y. Jen, “Ultralarge and thermally stable electro-optic activities

from supramolecular self-assembled molecular glasses,” J. Am. Chem. Soc. 129(3), 488–489 (2007).

7. Y. Wang, D. L. Frattarelli, A. Facchetti, E. Cariati, E. Tordin, R. Ugo, C. Zuccaccia, A. Macchioni, S. L.

Wegener, C. L. Stern, M. A. Ratner, and T. J. Marks, “Twisted π-Electron System Electro-Optic Chromophores.

Structural and Electronic Consequences of Relaxing Twist-Inducing Non-Bonded Repulsions,” J. Phys. Chem. C

112(21), 8005–8015 (2008).

8. E. C. Brown, T. J. Marks, and M. A. Ratner, “Nonlinear response properties of ultralarge hyperpolarizability

twisted π-system donor-acceptor chromophores. Dramatic environmental effects on response,” J. Phys. Chem. B

112(1), 44–50 (2008).

(C) 2010 OSA 29 March 2010 / Vol. 18, No. 7 / OPTICS EXPRESS 6779#122281 - $15.00 USD Received 5 Jan 2010; revised 26 Feb 2010; accepted 1 Mar 2010; published 17 Mar 2010

9. H. Kang, A. Facchetti, H. Jiang, E. Cariati, S. Righetto, R. Ugo, C. Zuccaccia, A. Macchioni, C. L. Stern, Z. Liu,

S.-T. Ho, E. C. Brown, M. A. Ratner, and T. J. Marks, “Ultralarge hyperpolarizability twisted π-electron system

electro-optic chromophores: synthesis, solid-state and solution-phase structural characteristics, electronic

structures, linear and nonlinear optical properties, and computational studies,” J. Am. Chem. Soc. 129(11),

3267–3286 (2007).

10. G. Xu, Z. Liu, J. Ma, B. Liu, S. T. Ho, L. Wang, P. Zhu, T. J. Marks, J. Luo, and A.-K. Jen, “Organic electro-

optic modulator using transparent conducting oxides as electrodes,” Opt. Express 13(19), 7380–7385 (2005).

11. Y. Enami, C. T. Derose, D. Mathine, C. Loychik, C. Greenlee, R. A. Norwood, T.-D. Kim, J. Luo, Y. Tian, A.

K.-Y. Jen, and N. Peyghambarian, “Hybrid polymer/sol–gel waveguide modulators with exceptionally large

electro–optic coefficients,” Nat. Photonics 1(3), 180–185 (2007).

12. Y. Enami, D. Mathine, C. T. DeRose, R. A. Norwood, J. Luo, A. K.-Y. Jen, and N. Peyghambarian, “Hybrid

cross-linkable polymer/sol-gel waveguide modulators with 0.65V half wave voltage at 1550nm,” Appl. Phys.

Lett. 91(9), 093505 (2007).

13. D. Jin, D. Huang, B. Chen, H. Chen, L. Zheng, A. Barklund, G. Yu, E. Miller, M. Moolayil, B. Li, and R. Dinu,

“Low half-wave voltage modulators using nonlinear optical polymers,” Proc. SPIE 6653, 66530Q (2007).

14. R. Dinu, D. Jin, D. Huang, M. K. Koening, A. M. Barklund, and Y. Fang, “T. C. Parker Z. Shi, J. Luo and A. K.-

Y. Jen, “Low-voltage electro-optic polymer modulators,” Proc. SPIE 6243, 62430G (2006).

15. Y. Shi, W. Lin, D. J. Olson, J. H. Bechtel, H. Zhang, W. H. Steier, C. Zhang, and L. R. Dalton, “Electro-optic

polymer modulators with 0.8 V half-wave voltage,” Appl. Phys. Lett. 77(1), 1–3 (2000).

16. R. Harold Fetterman and William H. Steier, “High-Speed Polymer Optical Modulators and Their Applications,”

in High-speed photonic devices, by Nadir Dagli, Taylor & Francis (2003)

17. J. K. Luo, and H. Thomas, “Transport properties of indium tin oxide/p-InP structures,” Appl. Phys. Lett. 62(7),

705 (1993).

18. M. J. Tsai, A. L. Fahrenbruch, and R. H. Bube, “Sputtered oxide/indium phosphide junctions and indium

phosphide surfaces,” J. Appl. Phys. 51(5), 2696 (1980).

19. M. Yan, M. Lane, C. R. Kannewurf, and R. P. H. Chang, “Highly conductive epitaxial CdO2 thin films prepared

by pulsed laser depostion,” Appl. Phys. Lett. 78(16), 2342 (2001).

20. X. Wang, S. Yang, J. Wang, M. Li, X. Jiang, G. Du, X. Liu, and R. P. H. Chang, “Structural and optical

properties of ZnO films by plasma-assisted MOCVD,” Opt. Quantum Electron. 34(9), 883–891 (2002).

21. M. Yan, Y. Koide, J. R. Babcock, P. R. Markworth, J. A. Belot, T. J. Marks, and R. P. H. Chang, “Selective-area

atomic layer epitaxy growth of ZnO features on soft lithography-patterned substrates,” Appl. Phys. Lett. 79(11),

1709 (2001).

22. R. Asahi, A. Wang, J. R. Babcock, N. L. Edleman, A. W. Metz, M. A. Lane, V. P. Dravid, C. R. Kannewurf, A.

J. Freeman, and T. J. Marks, “First-Principles Calculations for Understanding High Conductivity and Optical

Transparency in InxCd1-xO Films,” Thin Solid Films 411(1), 101–105 (2002).

23. A. J. Freeman, K. R. Poeppelmeier, T. O. Mason, R. P. H. Chang, and T. J. Marks, “Chemical and Thin Film

Strategies for new Transparent Conducting Oxides,” Mats. Res. Soc. Bull. 25, 45–51 (2000).

24. L. Wang, M. H. Yoon, G. Lu, Y. Yang, A. Facchetti, and T. J. Marks, “High-performance transparent inorganic-

organic hybrid thin-film n-type transistors,” Nat. Mater. 5(11), 893–900 (2006).

25. R. M. Corless, G. H. Gonnet, D. E. G. Hare, D. J. Jeffrey, and D. E. Knuth, “On the Lambert W function,” Adv.

Comput. Math. 5(1), 329–359 (1996).

26. COMSOL Multiphysics 3.5, COMSOL AB.

27. Lumerical Mode Solutions 3.01 trial, Lumerical Solutions Inc.

28. S. Y. Wang, and S. H. Lin, “High speed III-V electrooptic waveguide modulators at λ = 1.3 µm,” J. Lightwave

Technol. 6(6), 758–771 (1988).

29. High Frequency Structure Simulator 9.1, Ansoft Corp.

1. Introduction

Many aerospace and telecommunications applications require greatly increased bandwidths

and lower operating power for high-speed data transmission and analysis, which could, in

principle, be enabled by lower power, higher speed electro-optic (EO) modulators than are

currently available. One important application area requiring high-speed EO modulators with

substantially lower driving power is RF photonics, which promises replacement of electrical

RF transmission lines with much lighter weight, more power-efficient RF optical links. RF

photonics, when fully developed, will have many application spaces, including antenna

remoting, antenna beam formation, signal synthesis, frequency conversion and

channelization, as well as radar and communications [1]. Another important application area

that would benefit from ultra-low power EO modulators is efficient optical interconnects,

which would enable next-generation microprocessors [2].

A critical performance requirement in RF photonics and optical interconnect design is that

the operating power requirements of the enabling high-speed modulators be as low as


possible so that minimum power semiconductor lasers can achieve efficient, near-unity RF-

to-optical power conversion. For example, commercial LiNbO3 modulators have switching

voltages (also called the π-phase-shift or half-wave voltage, Vπ) of 5V, an active length of L =

2cm, and an electrical terminal impedance of Z = 50Ω [3]. This means that the electrical

power required to drive the modulator, given by P = Vπ2/Z, is ~500mW, which is excessive in

terms of electrical-to-optical signal power conversion since the semiconductor laser powers

used in fiber optic communication are typically less than 10mW. Note that the required

electrical power scales as (Vπ)2, so that there is a great advantage to reducing Vπ. For example,

a 5.0V→ 0.5V reduction in Vπ reduces LiNbO3 modulator driving power requirements 100-

fold to 5mW, and if the total optical insertion loss is held below 6dB, a nearly one-to-one

electrical-to-optical signal power conversion can be achieved with a 10mW semiconductor

laser. Recently, there have been substantial advances in polymeric EO materials with r33

coefficients reaching the impressive 200pm/V regime–more than 5x larger than that of

LiNbO3 [4–9]. EO modulators using these organic EO materials and conventional device

structures [e.g., Fig. 1(a)] have been reported with in-device r33s from 140 - 170pm/V [11,12].

Although these devices have reached Vπ = 0.66 – 1.0 V, the great potential of these new

materials is not realized in conventional modulator designs, due to the large voltage drop

across the thick cladding layers required to avoid metal electrode-induced optical loss.

Typical interelectrode distances in these devices range from 8 - 15µm, and the corresponding

active electrode lengths Les are necessarily large, 2.4 - 3.5cm, since Vπ scales inversely with

Le. However, such long active lengths limit the electrical bandwidth by increasing the RF-

optical velocity mismatch and RF loss in the active region. Therefore, achieving sub-1V EO

modulators with large modulation bandwidths and compact device lengths (sub-1cm) presents

a daunting challenge for conventional modulator designs.

Here we report an alternative approach which builds on the attractions of these organic

materials but drastically modifies the modulator design for maximum performance. This

includes replacing the thick cladding layers with non-metallic transparent conducting oxide

(TCO) bridge electrodes, or inserting thin TCO layers between the cladding and core layers.

In both strategies, the TCO acts as a “bridge” to conduct the driving voltage from the metal

transmission line directly to the EO layer. The interelectrode distance is then reduced from

the thickness sum of the top cladding layer + the active layer + the bottom cladding layer, to

the thickness of the active layer only.


Fig. 1. Evolution of a conventional organic EO modulator structure to a TCO electrode-based

organic EO modulator design using a side conduction geometry. (a) Conventional organic EO

modulator structure having two thick waveguide cladding layers that separate the metal

electrodes from the EO optical waveguide core to reduce metal-induced optical loss. (b) TCO-

based modulator having a top-down conduction geometry utilizing a TCO material as the

waveguide cladding layers to conduct the driving voltage directly from the metal electrodes to

the EO waveguide core. This geometry is limited by the high refractive index of typical TCO

materials versus that of organic EO materials. (c) TCO-based EO modulator design with a side

conduction geometry utilizing a pre-etched trench structure in the bottom cladding layer to

form an effective optical waveguide, and two thin TCO layers acting as bridge electrodes to

laterally deliver the switching voltage to the waveguiding EO core from metal side electrodes.

In previous work, we briefly reported an early modulator design using side-conducting

TCO electrodes, a top cladding arrangement (a horizontal multi-mode “effective”

waveguide), and a low-r33 organic EO material [10]. We also estimated the modulation

bandwidth. In the present contribution, we now report: 1) a new, more easily fabricated

“inverted rib waveguide” design with better polymer compatibility, 2) incorporation of a

high-response organic EO material (AJCKL1) with a larger r33 and greater thermal stability

[4], 3) a comprehensive analytical model to optimize Vπ , the optical insertion loss, and the

modulation bandwidth based on full wave numerical simulation. 4) a new buried waveguide

structure with a side conduction geometry which minimizes RF loss and velocity mismatch,

yet maintains good electrical-optical confinement. The sum of these results shows that TCO

electrode-based modulators combining current-generation EO polymers (r33 = 100 -

200pm/V) offer Vπ = 0.5V, high operation frequency potential (40-100GHz), and compact

dimensions (5mm active length). Such devices should be ideal for RF photonics applications,

with the compact sizes capable of on-chip integration with semiconductor lasers.

2. Voltage-size figure-of-merit enhancement using TCO modulator electrodes

The voltage-size figure-of-merit for an EO modulator is given by the product of the half-wave

voltage Vπ and the length L of the device [Eq. (1)], where λ is the optical wavelength, n is the

3

el sepdV L

n rπ

λ −=Γ

(1)

EO polymer refractive index, Г is the overlapping factor of the light field within the EO

region, and r is the polymer effective EO coefficient. In a single-waveguide polarization

interference geometry, r = (r33 – r13) = (2/3)r33 (assuming r33 ~1/3r33), while in a dual-

waveguide “push-pull” Mach-Zehnder modulation geometry, r is effectively 2r33 [11]. The r33

coefficient defines the refractive index change in the modulating electric field direction. In

many applications, it is equally meaningful to specify the power-size figure-of-merit given

by: PπL2 = (Vπ

2L

2)/Z0, where Z0 is the transmission line impedance that is typically 50Ω. We

see from Eq. (1) that Vπ is proportional to the electrode-electrode distance del-sep. For the

conventional modulator of Fig. 1(a), del-sep is much larger than the EO layer thickness dcor due

to the necessity of having two thick cladding layers (top layer thickness = dtcl and bottom


layer thickness = dbcl) required to separate the metal electrodes from the EO active region to

suppress metal-induced optical loss (i.e., del-sep = dcor + dtcl + dbcl). Typical organic modulators

operating at 1550nm have del-sep ranging from 8 −15µm with a typical value of 12 µm and dcor

≈1.5µm [11–15]. However, note that if the cladding layers have sufficient optical

transparency and electrical conductivity, they can introduce the switching voltage directly

from the metal electrode to the EO layer, thus greatly reducing Vπ. However, a limitation is

that typical TCO materials have high refractive indices at telecommunication wavelengths,

nTCO > 1.7 (e.g., nIn2O3 = 1.75 - 2.05, depending on the doping density), significantly greater

than those of typical organic EO materials, nEO = 1.4 - 1.7 [4–6], thus rendering TCOs

unsuitable cladding materials. Furthermore, even if TCOs could be used as cladding, their

optical loss may be too large for efficient waveguiding. To address this limitation, we employ

the approach of Fig. 1(c). Here two thin TCO layers conduct the voltage from metal side

electrodes to the top and bottom parts of the EO waveguide core, thereby reducing del-sep from

del-sep = dcor + dtcl + dbcl to del-sep = dcor. Table 1 below compares the voltage-size figure-of-

merit enhancement provided by the new electrode geometry. A 1cm-long device with a

conventional push-pull metal electrode design requires an EO material having a large r33 =

600pm/V to achieve Vπ = 0.5V. However, if the EO layer is directly modulated by transparent

electrodes as in Fig. 1(c), the r33 required for Vπ = 0.5V is only 75pm/V. Furthermore, for sub-

milliwatt operation (e.g., 200µW for a 1 cm active length push-pull design), a conventional

modulator structure requires an EO material having a currently unattainable r33 = 3000pm/V,

while the TCO-based modulator requires only r33 = 375pm/V, which is currently possible

[4–9].

Table 1. Projected enhancement of EO modulator power-size figure-of-merit using a

TCO electrode-based device versus a conventional structure. A TCO-based structure can

enhance an organic EO modulator power-size figure-of-merit by 10x-100x versus a

conventional device structure using the same EO material. We assume nEO = 1.6 and ГEO

= 80%.

Vπ

(Assuming 1cm active

length with push-pull

design)

Required Power

(PRF = V2/50Ω)

Required r33 with Metal

Electrodes

(del-sep = 12µm)

Required r33 with

Transparent Electrodes

(del-sep = 1.5µm)

5V 500mW 60 pm/V 7.5 pm/V 1V 200mW 300 pm/V 37.5 pm/V

0.5V 5mW 600 pm/V 75 pm/V 0.25V 1.25mW 1200 pm/V 150 pm/V 0.1V 200µW 3000 pm/V 375 pm/V

3. Modulator fabrication and evaluation results

New Device Structure. To demonstrate the low VπL potential of the present TCO electrode-

based EO polymer modulators, a proof-of-concept device was fabricated. We ultilize

AJCKL1 with a larger r33 and better orientational thermal stability than that used in earlier

work [10]. A schematic and SEM cross-section image of the straight-channel phase

modulator are shown in Fig. 2(a); Fig. 2(b) shows the waveforms of the applied switching

voltage at 1KHz and the intensity of the modulated light beam. Note the differences from the

first-generation design of Fig. 2(c) [10]. The new structure in Fig. 2(a) has far better

compatibility with the EO material because the optical mode confinement in horizontal

direction is realized with an inverted rib waveguide, formed by pre-etching a trench in the

bottom cladding layer (SiO2) before spin-coating and poling the EO polymer layer. This

strategy avoids undesirable thermal exposure in the post-poling process–fabrication of the top

cladding layer (NOA74 in Fig. 2(c) by photolithography and RIE etching as in previous work

[10]. The deposition of the top TCO electrode (In2O3) after the EO poling process can be

carried out at room temperature.


Fig. 2. (a) Inverted rib waveguide structure used for the present demonstration of a TCO

electrode-based organic EO modulator; and SEM cross-section image of the fabricated device;

(b) Measured waveforms of the switching voltage and the detected output light intensity. The

measurement was made at λ = 1.3µm, with a driving voltage frequency of 1kHz. (c) Earlier

device structure from ref. [10]. (d) Mode pattern for TE / TM mode and a photo of the mode

taken with a CCD camera. In the simulation, we assume nEO = 1.68 and nTCO = 2.0. The new

structure has higher ГEO ≥ 95% than in previous work because the top cladding layer is now air

rather than NOA74. The optical mode overlapping factor in the top TCO layer, the EO layer

and the bottom TCO layer for both TM and TE mode are listed. The total TCO overlapping

factor is ГTCO = 0.78% for TM mode and ГTCO = 1.71% for TE mode.

TCO modulator fabrication. Process steps are shown in Fig. 3. After fabrication of the

inverted rib structure on a 3µm thick SiO2 layer on Si by standard photolithography (step 1), a

60nm In2O3 TCO layer was grown by Ion-Assisted Deposition (IAD) at room temperature

[23] to form the bottom TCO bridge electrode (step2). A 90nm SiO2 layer was then grown on

top of the TCO layer as protective layer to reduce EO polymer breakdown during electric

field poling (step 3). A 150nm gold layer was then thermally evaporated/patterned by shadow

masking on the side of the bottom TCO bridge electrode to act as the bottom contact (step 4).

Next, a 1.5µm EO polymer layer was spin-coated onto the substrate. The AJ-CKL1 EO

polymer was formulated by doping 30 wt% of chromophore AJY02 into a low-loss, high-Tg

amorphous polycarbonate (APC) host. Thus, 32.0 mg of APC and 13.7 mg of AJY02 was

dissolved in 770 mg of dibromomethane, and the mixture shaken for 3 h to obtain a

homogeneous solution. This solution was then filtered through a 0.2 µm pore size PTFE filter

and spin-coated onto the device substrate, followed by drying under vacuum at 80°C for 0.5

h. The film thickness was adjusted to ~1.5-1.8 µm using a 1100-1600 rpm spinning speed.

Under these processing conditions, the EO polymer filled the trench and formed the desired

inverted rib optical waveguide structure. The refractive indices of this polymer in its unpoled

form were measured with a MetriCon 2010 Prism Coupler and are estimated to be 1.693 (TE)

and 1.680 (TM) at 1300 nm; and 1.661 (TE) and 1.642 (TM) at 1550 nm(step 5). The EO

polymer fills the trench and forms the desired inverted rib optical waveguide structure. Next,

a solution of 20 wt% poly(4-vinylphenol) (PVP) in n-propanol was spin-coated on the top of

the EO layer to deposit a 1.5µm PVP film (step 6). The PVP/EO bilayer was then thoroughly

dried under vacuum at 80 °C. This PVP layer is used as a sacrificial buffer for poling, and is

essential to produce poled EO films with acceptable surface quality. We explored the

undoped version of PVP (resistivity ~1x1010Ω*m) and a PEDOT-PSS doped version

(resistivity ~1x106Ω*m), to see whether the doped PVP layer might enhance the poling

efficiency versus the undoped PVP layer. After the PVP layer deposition and drying, a 100nm

gold poling electrode was thermally evaporated onto the PVP layer (step7). The poling


voltage applied to the EO/PVP stack was ~250V-300V for the device with undoped PVP

layer and 150V – 250V for the device with a doped PVP layer. The film temperature was then

ramped from 60°C to 135°C at a rate of 10°C/min. The poling current increased during the

temperature ramp process. The maximum current observed during poling of the EO/PVP

stack was 5µA for undoped PVP (250V poling voltage), and 30µA for doped PVP (225V

poling voltage). Under the same poling profile, the maximum current during the poling of the

single EO layer is 170µA, which means that the PVP protective layer reduces the maximum

achievable poling current.

Fig. 3. Fabrication process for the side conduction TCO electrode-based EO modulator of

Fig. 1(c).

Once the maximum temperature is reached, the sample is slowly cooled to room

temperature before terminating the voltage (step 8). After the EO poling process, the gold

poling electrode was removed by wet etching and the PVP protective layer removed with

ethanol (step 9). A 60nm In2O3 TCO layer was then grown by IAD at room temperature on

top of the poled EO film to form the top TCO bridge electrode (step 10). Finally, a 150nm

gold layer is thermally evaporated/patterned on top of the TCO bridge electrode to form the

top metal contact (step11).

Measurement of VπL figure-of-merit. The EO phase shift was measured by converting the

phase modulation to intensity modulation using a cross-polarization interference setup: input

light from a 5mW 1310nm semiconductor laser was linearly polarized at + 45° to the

direction of the switching electric field. The light was coupled into the straight waveguide

using a 60x objective lens with a numerical aperture of 0.6. The output light from the

waveguide was collected by another 60x objective lens and passed through a polarization

analyzer oriented with an analyzed polarization at −45° to the direction of the switching

electric field. Phase modulation was converted to intensity modulation after the analyzer. The

intensity modulation is detected by a photo-detector and recorded with an oscilloscope.

For a number of devices fabricated, the observed voltage-size figure-of-merit ranged from

0.6V-cm to 0.9V-cm, after physically reasonable conversion of the directly measured VπL to

that for a push-pull geometry. We summarize details of the measurement results in Table 2.

As discussed above, for the sacrificial poling protective layer (PVP), we explored both un-

doped and doped (conductive) PVPs. For each PVP formulation, we also varied the poling

voltage. From Table 2 note that the device with the conductive PVP layer and higher poling


voltage provides the smallest VπL product voltage-size figure-of-merit, 0.6V-cm, which

corresponds to a very low power-size figure-of-merit of 7.2mW-cm2–very close to the desired

5mW driving power in a 1.0cm long device. In our previous work, we reported Vπ = 2.8V

with in an 8mm straight waveguide device or VπL = 0.75V-cm (converting to a push-pull

value), with an inter-electrode distance of 1.5µm, based on a previous generation EO polymer

(AJLS8/APC). While the best result of the present work is 0.56V-cm with an inter-electrode

distance of 2µm (0.5µm larger than previously due to the trench in SiO2 layer), there is an

effective 45% reduction in VπL (0.56V-cm/2µm versus 0.75V-cm/1.5µm).

Using cut-back methods, the measured waveguide loss of the present devices is ~4dB/mm

for the TM mode and 8dB/mm for the TE mode, with a coupling loss of 9dB. In this work,

In2O3 (α ~1000/cm, σ = 70S/cm) is used for both the top and bottom TCO electrodes. Since

the bottom TCO layer has much larger mode overlapping factor ГTTCO than the top TCO layer

(from Fig. 2, ГBTCO = 0.71% and ГTTCO = 0.07% for the TM mode), the optical loss caused by

the bottom In2O3 electrode is 10log(e-1000/cm*0.71%*L

)/L = 3.1dB/mm and the optical loss caused

by the top In2O3 is 0.32dB/mm, with the total TCO induced loss = 3.4dB/mm, close to the

measured value. Therefore, in the new modulator structure, the waveguide loss mainly comes

from the bottom TCO layer. Since the TCO induced optical loss is proportional to the product

of αTCO and ГTCO, it can be reduced by engineering the TCO material to have a lower αTCO, or

by inserting thin buffer layers between the EO and TCO layers to reduce ГTCO. The first

method requires TCO materials development, and a systematic study of how to reduce the

αTCO while keeping a high σTCO during TCO deposition. This will be discussed in a later

publication. In Section 3 we discuss the second strategy in detail.

Table 2. Detailed measurement results of TCO electrode-based organic EO modulators.

Device # Poling

protective

layer

Poling

Voltage Vπ L VπL

(converted to

push-pull

geometry)

1 Un-doped 200V 8.1V 3.0mm 0.81V-CM 2 Un-doped 250V 5.3V 3.8mm 0.67V-CM 3 Doped 150V 9.3V 2.0mm 0.62V-CM 4 Doped 225V 11.2V 1.5mm 0.56V-CM

4.Comprehensive modeling work for a new high speed modulator structures

In our previous work, we estimated the modulation bandwidth of the first generation TCO

electrode-based organic EO modulator shown in Fig. 2(c). Although the structure is easy to

fabricate and demonstrate operation at low frequencies, it is by no means optimized,

especially for high speed operation, because: 1) horizontally it is a multimode optical

waveguide since the lateral mode confinement is through an “effective” top cladding layer, 2)

the overlapping area of the two TCO bridge electrodes must be as large as 4µm to achieve

good electrical to optical mode overlapping, and this will cause a large RF loss in the TCO-

EO active region, 3) the TCO layer thickness was deliberately kept thin to avoid refractive

index mismatch and also to reduce the optical loss in the TCO layers. In that work, we used a

50nm thick bottom TCO electrode and a 20nm thick top TCO electrode. In the present study,

we systematically analyze the interplay of half-wave voltage Vπ, optical insertion loss, and

modulation bandwidth. The effect of the TCO layers on the optical mode will be discussed

based on a numerical simulation. We then optimize the structure to achieve better optical

confinement in the EO waveguide core and lower optical mode overlapping with in the TCO

layers. We also propose a new “buried waveguide” structure for high speed operation. The

effect of TCO bridge electrodes on the RF loss of the EO active region, or the “TCO loading

effects”, will be discussed in detail. Finally the modulation bandwidth will be predicted based

on the numerical simulation results.


The optical waveguide without the two TCO electrodes is shown in Fig. 4(a). The

refractive index of the EO material nEO is assumed to be 1.6 which is typical for organic EO

polymers. To efficiently confine the optical mode, the low index polymer CYTOP with n =

1.35 (ε = 2.1) was chosen as the material for the side and bottom cladding layers. The

thickness and width of the waveguide core are: dcor = 1.3µm and wcor = 1.2µm, so that it is a

single mode waveguide in both vertical and horizontal directions. The calculated optical

energy confinement factor in the waveguide core ГEO = 83%. If r33 = 150pm/V and a dual

drive structure are assumed (which means the effective EO coefficient r = 2r33 = 300pm/V),

then VπL = (λdel-sep)/(n3rГEO) = 0.205V-cm for λ = 1.55µm, and 0.173V-cm for λ = 1.31µm,

and an L = 5mm long active length yields Vπ = 0.41V for λ = 1.55µm operation and 0.34V for

λ = 1.31µm operation. At this point we assume that del-sep = dcor which means that the two

TCO electrodes directly contact the waveguide core. However, since the TCO layers will

cause optical loss in the waveguide due to free carrier absorption, the design of the two

electrodes is crucial to the overall device performance. Since the optical insertion loss is

proportional to the product of αTCO and ГTCO, an effective way to reduce the optical insertion

loss is by reducing ГTCO. Later we show that, by inserting thin buffer layers between the TCO

electrodes and the EO waveguide core, ГTCO can be reduced significantly, and this strategy

allows the TCO electrodes to have much higher electrical conductivity without significantly

increasing the VπL product. A higher TCO electrode electrical conductivity will greatly

benefit the modulation bandwidth by reducing both the RF loss in the TCO loaded EO active

region and the horizontal voltage drop along the TCO electrodes.

Fig. 4. (a) Buried waveguide structure and the resulting optical mode pattern. The EO

waveguide core (nEO = 1.6) is buried within two side cladding layers (nscl = 1.35) and a bottom

cladding layer (nbcl = 1.35). dcor = 1.3µm and Wcor = 1.2µm . The optical waveguide is single

mode in both the horizontal and vertical directions. The effective refractive index neff = 1.467

and the TM optical mode confinement factor in the waveguide core ГEO = 83% (b) buried

waveguide with two TCO electrodes (nTCO = 1.9) and buffer layers, and the optical mode

pattern. The bottom buffer layer is 300nm thick and the top buffer layer is 100nm thick. The

arrows show the direction of the E-component of the optical mode.

To determine the required TCO electrode dimensions, we must first analyze the optical

loss requirements of an EO modulator. The total loss of an optical waveguide device with

TCO layers is given by: Iout/Iin = TTCO × TMet × Toth, where TTC is the optical power

transmission coefficient accounting for the optical loss caused by the TCO layer alone, which

can be further described by: TTCO = exp(-αTCoptΓTCL), where L is the length of TCO layer in the

device (it is also the modulator interaction length) and ΓTCO is the percentage of optical mode

energy overlapping with the TCO layer (the TCO optical-mode overlapping factor). TMet is

the transmission coefficient accounting for the optical loss due to the optical mode touching

the metal transmission line on both sides and is given by TMet = exp(-αMetL). As mentioned

above, for an optimal design, we can let TMet = TTCO. A typical commercial LiNbO3 EO

modulator has a device optical insertion loss of less than 6 dB (< 75% loss in optical power).


The typical fiber coupling loss at the input and output ports can typically be lower than 30%

(1.5dB) per port, yielding a total coupling loss of less than 50% (3dB). Assuming that other

propagation losses, including EO material absorption loss, total to be less than 20% (1dB),

Toth will be no less than (1 − 0.3) × (1 − 0.3) × (1 − 0.2) ≈0.4 (4dB). To achieve a similar total

device insertion loss of 6 dB for our modulator design, it is desirable to keep the optical

propagation loss due to the TCO and metal to be less than 40%, i.e., keep (TTCO × TMet) > 0.6

(less than 2.2dB) or TTCO > 0.775 (less than 1.1dB) assuming TTCO = TMet, so that (Toth × TTCO

× TMet) will be greater than 0.25 (< 75% loss or < 6 dB total device insertion loss). For here

and all examples given below, we assume an RF-optical interaction length L = 0.5 cm, so that

the optical transmission TTCO will be > 0.8 if αTCoptΓTC < ln(0.8)/L = 0.22/L (for L in cm)

which requires αTCopt ΓTCO < 0.44/cm when L = 0.5cm. Table 3 summarizes the optical loss

caused by each part of the TCO electrode-based EO modulator.

Table 3. Details of the optical insertion loss caused by each modulator component

Total Insertion loss: 6dB Coupling loss (2 ports) 3dB

Metal induced loss 1dB TCO layer induced loss 1dB

Other materials loss 1dB

Regarding TCO electrode composition, materials such as tin-doped indium oxide (ITO)

are used widely in flat panel displays. While ITO is excellent for visible wavelength

applications, it is not suitable for the 1550nm fiber-optic telecommunication wavelengths due

to the high IR optical absorption. Since the modulators of interest are intended to operate at

1550nm, TCOs with low optical absorption in this region are essential. For such applications,

TCOs such as undoped In2O3, ZnO, or CdO are more suitable due to their low optical

absorption at 1300-1550nm [17–23], as given by the loss coefficient αTCO. Besides the low

loss requirement, an electrode TCO material must have a sufficiently high electrical

conductivity, σTCO, to drive the modulator at high speed. While doping TCO materials with

additional carriers increases σTCO, it also increases free carrier absorption at longer

wavelengths, increasing αTCO [17–24]. The electrical conductivity to optical absorption

coefficient ratio, FTCO = σTCO/αTCO, is an intrinsic materials property at a given wavelength

and an important TCO modulator figure-of-merit. As discussed before, the acceptable αTCO is

inversely proportional to ΓTCO through the relationship αTCoptΓTCO < 0.44 when L = 0.5cm,

while σTCO is proportional to αTCO when a certain FTCO is assumed. Therefore, the optical

mode overlapping factor ΓTCO must be as small as possible to obtain as large a σTCO as

possible. This can be achieved by either making the two TCO layers very thin, or adding thin

buffer layers between the EO layer and the TCO layers. Table 4 shows the numerical

simulation results for the relationship between ΓTCO and the thicknesses of the TCO and

buffer layers. The results are given by COMSOL, which is an FEM method based mode

solver [26]. Here we define the optical mode overlapping factor in the bottom TCO layer to

be ΓBTCO and the optical mode overlapping factor in the top TCO layer to be ΓTTCO. The total

mode overlapping factor in the TCO layers is ΓTCO = ΓBTCO + ΓTTCO. Note that if the TCO

layers directly contact the EO layer, ΓTCO is 2.9% and the corresponding αTCopt = 15 /cm.

ΓBTCO is larger than ΓTTCO because the refractive index top cladding layer (air) is smaller than

the bottom cladding layer (CYTOP), and the optical mode is shifted towards the CYTOP side.

However if there is a 300nm thick bottom buffer layer and 100nm top buffer layer in between

the bottom TCO layer and the EO layer, ΓTCO will be reduced to 0.77%, and the

corresponding αTCopt becomes 57 /cm . Here we make dBBuff > dTBuff to assure that ΓBTCO ≈ΓTTCO

(i.e. separate bottom TCO layer away from the EO layer further than the top TCO layer). The

thickness of the two TCO layers TTCO is set here to 100nm. Assuming FTCO = 1S, the

corresponding σTCO = 57S/cm. Note that the 400nm thick buffer layer will only increase the

Vπ to 0.54V for λ = 1.55µm operation and to 0.44V for λ = 1.31µm operation. Later we will


show that higher TCO electrode electrical conductivity will greatly reduce the RF loss and the

horizontal TCO-RC voltage drop along the two TCO electrodes. Therefore we can choose

dBbuff = 300 nm and dTbuff = 100nm. In Table 4 we also list the overlap integral ηoverlap between

the TM mode of the structure with TCO layers, and the TM mode of the structure without

TCO layers, found by a commercial mode solver [27]. Here ηoverlap is defined as

* *

1 2 2 1

* *

1 1 2 2

( )( ) 1[ ]

( )overlap

E H d S E H d SRe

E H d S Re E H d S

η

→ →→ → → →

→ →→ → → →

× ×=

× ×

∫ ∫

∫ ∫

i i

i i

(2)

Note in Table 4 that the values of ηoverlap are all close to 100% which means that the

optical loss in the optical waveguide caused by the mode mismatch between the section

without TCO layers and the TCO loaded section can be ignored.

Table 4. Relationship between ГTCO and the thicknesses of buffer layers. The thickness of

the TCO layer TTCO = 100nm, and here we assume λ = 1.55µm. The data is for TM mode.

dBbuff (nm)

dTbuff del-sep ГBTCO ГTTCO ГTCO αTCopt ηOverlap

0 0 1.3µm 2.39% 0.50% 2.90% 15/cm 100% 100 0 1.4µm 1.24% 0.58% 1.82% 24/cm 97.6% 200 0 1.5µm 0.69% 0.64% 1.33% 33/cm 97.6% 300 0 1.6µm 0.39% 0.67% 1.06% 42/cm 97.3% 300 100nm 1.7µm 0.40% 0.37% 0.77% 57/cm 96.1%

In Fig. 5(b), we show the values of neff and ngopt at different optical wavelengths for the

modulator structure with dBbuff = 300 nm and dTbuff = 100nm. Here ngopt = nopt – λ0dnopt/dλ. The

group index optical waveguide ngopt is found to be 1.642 at λ = 1.31µm and 1.643 at λ =

1.55µm–slightly higher than neff. Later in the RF simulation, ngopt will be used to calculate the

velocity mis-match. Figure 5(c) shows the relationship between ГEO, ГTCO, and λ. Note that

ГTCO at λ = 1.31µm is smaller than ГTCO at λ = 1.55µm and therefore will allow the TCO

layers to have a larger αTCO. Figure 5(d) shows the relationship between αMet (calculated from

the extinction coefficient к) and Wgap. As discussed above, the metal-induced optical insertion

loss is required to be <1dB. This requires that we find a Wgap at which TMet = exp(-αMetL) >

0.775, or αMet < 0.44/cm when L = 5mm. In order to minimize Wgap, we first choose a large

refractive index difference between the EO waveguide core (nEO = 1.6) and the side cladding

layer (nscl = 1.35). This ensures a good horizontal mode confinement in the EO waveguide

core region. Secondly, as shown in Fig. 5(a), the two metal electrodes are located on the two

sides of the waveguide and the two TCO bridge electrodes extend to the top and bottom of the

EO waveguide core region. The modulator will work in TM mode since its E-component is

parallel to the RF electric field provided by the two TCO bridge electrodes. This arrangement

also helps minimize Wgap because the TM mode has a better horizontal confinement in the EO

waveguide core, compared with the TE mode case. In the numerical simulation, αMet is found

through the extinction coefficient к which is the imaginary part of the complex effective

refractive index of a mode (nc = n + iк) using the relationship: к = αMetλ/(4π). We see from

Fig. 5(d) that when Wgap > 1.6µm, αMet is small enough (below 0.44/cm for both λ = 1.31µm

and λ = 1.55µm).


Fig. 5. (a) Buried waveguide structure with two TCO electrodes and buffer layers. Two side

copper electrodes are also included to find the metal induced optical loss coefficient αMet. nscl =

nbcl = 1.35. dcor = 1.3µm, Wcor = 1.2µm, dBbuff = 300nm, dTbuff = 100nm, dTCO = 100nm. (b)

Effective refractive index and group index of the optical waveguide for TM mode c) The

optical mode confinement factor in the EO waveguide core ГEO and the optical mode

overlapping factor in the two TCO layers ГTCO = ГBTCO + ГTTCO, under different wavelength. d)

Metal-induced optical loss coefficient αMet versus the width of the gap Wgap.

Before going into the details of the RF transmission line design, we first analyze the

theoretical upper limit of the acceptable values of the microwave attenuation coefficient αRF

and TCO-RC voltage drop coefficient rRC. Figure 6 shows the standard RF model for the TCO

electrode-based EO modulator. An RF source launches the RF wave into the active region (a

TCO bridge loaded transmission line) through a standard Zs = 50Ω feeding transmission line.

A ZL = 50Ω termination is assumed at the end of the active region.

Fig. 6. Transmission line model for the TCO electrode-based EO modulator. The entrance

point of the TCO loaded EO active region is defined as x = 0. The termination point is defined

as x = L.

Assuming the voltage applied to the feeding transmission line is Vappl, then the

instantaneous RF voltage seen by the optical packet along the active region is given by:

(2 )

2( , ) [ ]

1

gopt goptRF RF

RF

x xj j

v vx L zEO

eff appl LL

metal L S

V TV x V e e e e

V e

ω ωγ γ

γω − − −−

= • • • + Γ−Γ Γ

(3)

Here γRF = αRF + jβRF is the complex propagation constant of the RF wave and ω = 2πf is

the angular frequency. αRF is the microwave attenuation coefficient which accounts for the RF

loss in the TCO loaded active region. βRF is the propagation constant which determines the


traveling speed of RF wave in the active region. RF reflection often happens when the

characteristic impedance of the active region Zm is not equal to Zs and ZL, and this is

accounted for by Гs = (Zs-Zm)/ (Zs + Zm), which is the RF reflection coefficient at the entrance

of the active region, and ГL = (ZL-Zm)/(ZL + Zm), which is the RF reflection coefficient at the

termination of the active region. T = 1- Гs is defined as the RF transmission coefficient at the

entrance of the active region. Because the TCO electrodes are often thin layers with limited

conductivity, the voltage they conduct to the EO waveguide core will drop below the voltage

on the metal transmission line, due to the TCO-RC loading effect, especially at the high

frequency. This fact is accounted for by adding an RC voltage drop coefficient: rRC =

VEO/Vmetal. The average switching voltage applied to the optical packet after it leaves the

active region, is found by integrating Veff(x,ω) along 0 to L:

2

0

1( ) (1/ )Re( ( , ) ) Re[ ( )]

FW FW RFLL L

L

av eff appl RC REF L

FW BW

e e eV L V x f dx V r r

L L

γγ γ

ωγ γ

−− −− −= = • +Γ∫ (4)

Here γFW and γBW are given by γFW = αRF + j (ω/c)(nRF-ngopt) and γBW = αRF + j (ω/c)(nRF +

ngopt).

Equation (4) gives the complete analytical model to predict how the effects of the applied

voltage will change with increased frequency. Note that our model is only slightly different

from the model given in [28] because here we add the TCO-RC drop factor rRC into the model

and also we define the entrance of the transmission line as x = 0.

The challenging part of designing a high-speed (3dB optical bandwidth>40GHz) TCO

based organic EO modulator is to manage the RF propagation loss along the TCO loaded EO

active region (denoted by αRF) and the voltage drop along the TCO bridge electrodes (denoted

by rRC). To find the theoretical upper limit of the RF loss, we can assume perfect impedance

matching between the feeding transmission line and the active region, and a perfect velocity

match between the RF wave and the optical wave, by setting Zm = Zs = ZL. and βRF = ω/vRF =

ω/vgopt (which means ngRF = ngopt). Later we will show that this can be achieved by carefully

selecting the dielectric materials and tuning the transmission line dimensions. Then, (ω/vgRF)

− βgRF = 0 and Eq. (4) becomes:

( )

(1 )( ) ( )

( )

RF f L

av eff appl RC

RF

eV f V r f

f L

α

α

−

−

−= • • (5)

We can see that now the averaged effective switching voltage Vav-eff(f) is determined by

two factors: one is the TCO-RC voltage drop factor rRC(f), which is a function of frequency f,

and the other is the RF decay factor (1 − e-x

)/x, in which x = αRF(f)L. The RF decay factor is

also a function of frequency f because the microwave attenuation coefficient αRF(f) will

increase with frequency f. The modulation bandwidth of an EO modulator can then be found

by solving for the frequency fBW at which the Vav-eff drops to Vappl/2 (optical 3dB bandwidth,

3dBo) or

( ) ( )(1 )

( ) 0.5( )

RF BWf Lav eff BW

RC BW

RF BW appl

V fer f

f L V

α

α

−−−

• = = (6)

In other words, if we have a targeted bandwidth fBW, the requirement for αRF(f)L is given

by:

0.5

1 , here ( )( )

x

RF BW

RC BW

e x x f Lr f

α−− = = , (7)

The solution of Eq. (7) is:

( )

0.5( ) ( )

( ) ( )0.5 0.5

RC BWr f

RC BW RC BW

RF BW

r f r fx f L W eα

−= = − + (8)


Here W(x) is the Lumbert W function [25]. Assuming rRC(fBW) = 1, or no RC voltage drop

along the TCO electrodes, then αRF(f)L = W(−2e−2

) + 2 = 1.6, or αRF(f) = 3.2 /cm if L = 0.5cm.

Here αRF is defined as the microwave attenuation coefficient of the electric field amplitude of

the RF wave, therefore 3.2/cm corresponds to 20log(e3.2L

)/L = 27.7dB/cm. For example, if the

target 3dB optical bandwidth is 40GHz, then the theoretical upper limit of αRF(40GHz) is

3.2/cm for a device with a 5mm long active length. If the TCO-RC voltage drop along the

TCO electrode factor rRC is taken into consideration, then the upper limit of αRF(fBW) will be

reduced to a lower value. Figure 7 shows the theoretical upper limit of αRF(fBW) under

different rRC values. It can been seen that if the frequency cutoff is already determined by the

TCO-RC voltage drop frequency cutoff factor (e.g., when rRC(f) ~0.5), there is little room for

the RF loss, and a small αRF(f) will push it to the cutoff (when the voltage drops to half). More

importantly, when L is increased from 5mm to 2cm, the theoretical upper limit of αRF when

rRC = 1 (no TCO-RC voltage drop effect) reduces by 4x to only 6.9dB/cm. This means the

room left for RF loss shrinks with increased active length.

In a TCO-enabled organic EO modulator, the RF loss comes from the metal transmission

line and the loading effect of the TCO electrodes in the EO active region. The metal

transmission line loss in the organic EO modulator structure is mainly caused by the skin-

effect of the metal electrodes and can be reduced [14,16]. Assuming the RF loss from the

metal transmission line to be as high as 7.7dB/cm at 40GHz, we still have 20dB/cm left for

TCO-induced RF loss. The actual RF loss in the TCO electrode loaded region depends on the

device structure and the TCO electrical conductivity. Later we will discuss the RF loss in

detail using a specific example. Note here that the active length L plays an important role in

the theoretical upper limit of the RF loss and the TCO-RC voltage drop. For example, in a

conventional organic EO modulator with a typical del-sep = 9µm (6x larger than a TCO-based

structure), the required active length L becomes 3cm, and the acceptable αRF(f) is reduced by

6x to 0.53 /cm, or 4.63dB/cm, which is only enough for the metal transmission line loss.

Fig. 7. Theoretical upper limit of the microwave attenuation coefficient αRF versus rRC under

different active lengths L. Here we assume perfect impedance match and velocity match.

Therefore, we see that the TCO electrode-based EO modulator has high speed potential

(f3dBo > 40GHz) because it can be made short (<5mm active length) while still achieving low

switching voltages (Vπ < 0.5V with dual drive) with currently available organic EO materials

(in device r33 = 150pm/V).

Now we give a specific example of an RF design based on full wave simulation to show

the effect of the two TCO electrodes on the device performance and how to optimize the

structures to achieve large modulation bandwidths. Figure 8 shows the proposed buried

waveguide with a coplanar slot transmission line and TCO side conduction geometry. In this

structure, the two parallel metallic plates to the left and right of the EO waveguide core form

a high-frequency RF transmission line. In the active region, the TCO material forms “bridge

electrodes” to transmit the voltage laterally from the metallic transmission line to the active

EO material region.


Fig. 8. High frequency structure design for a TCO-electrode based organic EO modulator. The

structure comprises a pair of high speed metallic transmission lines and two TCO electrodes

conducting the voltage from the metallic transmission line to the EO optical waveguide core.

In the numerical simulation, we use Wcor = 1.2µm, Wgap = 1.6µm, Wcopper = 250µm, dcor =

1.3µm, dTbuff = 100nm, dBbuff = 300nm, del-sep = dcopper = dcor + dBbuff + dTbuff = 1.7µm, dTCO =

100nm, dbcl = 3µm, dsub = 100µm, εEO = εTCO = 3, εscl = εbcl = εsub = 2.1. In practice, a low-k

substrate is required.

The structure is modeled using HFSS, which is a commercial finite element method

(FEM) solver [29]. The simulated electric field pattern of the RF mode in the TCO-EO core

region is shown in Fig. 8. It can be seen that in the EO waveguide core region, the RF electric

field is perpendicular to the TCO bridge electrodes and the electric field strength is uniform

across the entire EO waveguide core region. This ensures a good RF electric field - optical

mode overlapping in the EO waveguide core. Figure 9 shows the numerical simulation results

of the RF loss, the TCO-RC voltage drop factor rRC, the characteristic impedance Z0, and the

RF transmission line effective group index ngRF. Figure 9(a) and 9(b) show that the structure

is optimized so that Z0 is tuned to match 50Ω and the ngRF also matches ngopt (~1.64). This

means that the effects of the impedance mismatch and RF-optical velocity mismatch are

minimized. From Fig. 9(c) and 9(d) we can see that αRF and rRC depend on the electrical

conductivity of the TCO electrodes which is in turn dependent on the TCO material figure of

merit FTCO = σTCO/αTCO where αTCO = 57/cm is determined by the optical insertion loss


requirement discussed before. We explored the FTCO = 0S, 0.2S, 0.5S and 1S cases and the

corresponding σTCO = 0S/cm, 11.4S/cm, 28.5S/cm and 57S/cm, cases, respectively. The curve

with σTCO = 0S/cm represents the case in which only pure copper coplanar slot transmission

lines are present. Note that αRF in this case can be quite small. At 40GHz, the predicted αRF in

this situation is as low as 0.1328/cm or 1.15dB/cm. However, when the two TCO electrodes

are loaded in the active region, αRF will increase significantly. Note from Fig. 9(c) that when

σTCO = 11.4S/cm or FTCO = 0.2S, αRF at 40GHz becomes 1.0898/cm, or 9.466dB/cm. This is

because the RF wave interacts with the two TCO electrodes which have only finite electrical

conductivity. Although αRF is significantly higher than the value when there are only pure

metallic transmission lines (0.1328/cm or 1.15dB/cm), it is still below the theoretical upper

limit because in this case rRC = 0.81718 [Fig. 9(d)] and from Fig. 7 we find that the

corresponding theoretical upper limit of αRF at 40GHz is 2.15/cm or 18.68dB/cm. Note that

the structure is already optimized to achieve near impedance and velocity match. More

importantly, with TCO materials having larger FTCO values (0.5S or 1S), σTCO can be

increased to 28.5S/cm or 57S/cm. In these cases, at 40GHz, αRF will fall to 0.58/cm

(5.3dB/cm) for FTCO = 0.5S, or 0.37/cm (3.21dB/cm) for FTCO = 1S. And at the same time, rRC

increases to 0.956 for FTCO = 0.5S, and 0.984 for FTCO = 1.0S, which can be seen from

Fig. 9(d). This result shows the importance of the TCO figure of merit because a larger FTCO

will allow higher σTCO values, which in turn lead to lower αRF and higher rRC. From this trend,

note also the importance of short active length L and the buffer layers. For a device with a

short active length (~5mm), αTCO can be larger than the value in a conventional structure with

a 2cm long active length. Similarly, the buffer layers reduce the optical mode overlapping

factor with the TCO layers ГTCO and this in turn leads to larger αTCO (as shown in Table 4).

When FTCO is fixed, larger αTCO values mean larger σTCO.

Fig. 9. Simulation results for modulators having a TCO electrode loaded coplanar transmission

line (the active region), as shown in Fig. 8. (a) Characteristic impedance Z0 under different

σTCO. (b) Effective RF refractive index c) Microwave attenuation coefficient αRF d) TCO-RC

voltage drop coefficient rRC.


Note that the size of TCO overlapping area has a significant impact on the microwave

attenuation coefficient αRF. Figure 10 shows the value of αRF under different overlapping

widths Woverlap. When the TCO overlapping area width drops from 3.2µm to −0.8µm, αRF

drops from 0.445/cm or 3.86dB/cm to 0.195/cm or 1.69dB/cm. Here the σTCO is set to be

57S/cm (FTCO = 1S case). This is because the two overlapped TCO bridge electrodes act as a

loading capacitor with two series resistors. Increasing Woverlap will increase the loading

capacitance and the series resistance per unit length and therefore causes a larger RF loss

along the active region. To achieve good electrical-optical mode overlapping, the two TCO

electrodes must cover the entire EO waveguide core, meaning Woverlap ≥ Wcor Therefore, in the

design of the optical waveguide, we chose a low index material (CYTOP with n = 1.35) to be

the side cladding material, in order to minimize the width Wcor of the EO waveguide core. The

overall frequency response of the average applied voltage Vav is found by plugging the results

in Fig. 9(a)–9(d) into Eq. (4), and the final result of Vav is shown in Fig. 11. Note that 3dB

optical bandwidths (at which the average effective voltage drops to half the DC value) of

40GHz-100GHz can be achieved. Table 5 summarizes the complete device performance of

the proposed buried waveguide structure with coplanar metal transmission lines and the TCO

side conduction geometry.

Fig. 10. Microwave attenuation coefficient αRF versus the width of the TCO overlapping area

in the modulator structure shown.

Fig. 11. Overall frequency response of the average applied voltage found by plugging the

numerical results given in Fig. 9 into Eq. (4).


Table 5. Complete modeling result of the TCO based EO modulator.

Vπ L Required r33 Optical Insertion loss 3dB optical bandwidth ~0.5V 5.0mm 150 pm/V ≤6dB ≥40GHz

7. Conclusions

We have shown here that a transparent conducting oxide (TCO) electrode-based organic EO

modulator structure can be used to achieve substantially higher power-size performance than

conventional modulator designs. An optimized high-speed structure is proposed and its

performance carefully analyzed based on full wave numerical simulation. A compact device

(5mm long device) with 0.5V driving voltage and 40GHz – 100GHz optical bandwidth is

predicted. Note that a 0.5 V modulator operating at 40 Gbps in NRZ format has a switching

energy on the order of 63 fJ/bit. Experimentally, we have demonstrated a new EO modulator

structure which features an inverted ridge waveguide geometry offering a simpler fabrication

process, better thermal compatibility with the organic EO material, and a high-response EO

polymer active layer. Initial results demonstrate a power-size figure-of-merit of 7.2mW-cm2.

Compared with our previous experimental results, this represents an effective 45%

improvement.


Electro-optic modulator with exceptional power-size performance enabled by transparent conducting...

Documents

Transcript of Electro-optic modulator with exceptional power-size performance enabled by transparent conducting...